LinkSwitch-TN 设计指南en
Application Note AN-37 LinkSwitch-TN Family
https://www.wendangku.net/doc/1f295987.html,
April 2009
Design Guide
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Introduction
LinkSwitch-TN combines a high voltage power MOSFET switch with an ON/OFF controller in one device. It is completely self-powered from the DRAIN pin, has a jittered switching frequency for low EMI and is fully fault protected. Auto-restart limits device and circuit dissipation during overload and output short circuit (LNK304-306) while over temperature protection disables the internal MOSFET during thermal faults. The high thermal
shutdown threshold is ideal for applications where the ambient temperature is high while the large hysteresis protects the PCB and surrounding components from high average temperatures.LinkSwitch-TN is designed for any application where a non-isolated supply is required such as appliances (coffee machines, rice cookers, dishwashers, microwave ovens etc.), nightlights, emergency exit signs and LED drivers. LinkSwitch-TN can be con? gured in all common topologies to give a line or neutral referenced output and an inverted or non-inverted output
voltage – ideal for applications using triacs for AC load control. Using a switching power supply rather than a passive dropper (capacitive or resistive) gives a number of advantages, some of which are listed below.
Universal input – the same power supply/product can be used worldwide
High power density – smaller size, no μF’s of X class capaci-tance needed
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High ef? ciency – full load ef? ciencies >75% typical for 12 V output
Excellent line and load regulation High ef? ciency at light load – ON/OFF control maintains high ef? ciency even at light load Extremely energy ef? cient – input power <100 mW at no load Entirely manufacturable in SMD
More robust to drop test mechanical shock
Fully fault protected (overload, short circuit and thermal faults)Scalable – LinkSwitch-TN family allows the same basic design to be used from <50 mA to 360 mA
Scope
This application note is for engineers designing a non-isolated power supply using the LinkSwitch-TN family of devices. This document describes the design procedure for buck and buck-boost converters using the LinkSwitch-TN family of integrated off-line switchers. The objective of this document is to provide power supply engineers with guidelines in order to enable them to quickly build ef? cient and low cost buck or buck-boost converter based power supplies using low cost off-the-shelf inductors. Complete design equations are provided for the selection of the converter’s key components. Since the power MOSFET and controller are integrated into a single IC the design process is greatly simpli? ed, the circuit con? guration has few parts and no transformer is required. Therefore a quick start section is provided that allows off-the-shelf components to be selected for common output voltages and currents.
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Figure 1 (a). Basic Con? guration using LinkSwitch-TN in a Buck Converter. Figure 1 (b) Basic Con? guration using LinkSwitch-TN in a Buck-Boost Converter.
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In addition to this application note a design spreadsheet is available within the PIXls tool in the PI Expert design software suite. The reader may also ? nd the LinkSwitch-TN DAK
engineering prototype board useful as an example of a working supply. Further details of support tools and updates to this document can be found at https://www.wendangku.net/doc/1f295987.html,.
Quick Start
Readers wanting to start immediately can use the following information to quickly select the components for a new design, using Figure 1 and Tables 1 and 2 as references.
For AC input designs select the input stage (Table 9).
1.
Table 1. LinkSwitch-TN Circuit Con? gurations Using Directly Sensed Feedback.
Select the topology (Tables 1 and 2).
- If better than ±10% output regulation is required, then use optocoupler feedback with suitable reference.Select the LinkSwitch-TN device, L, R FB or V Z , R BIAS , C FB , R Z
and the reverse recovery time for D FW (Table 4: Buck, Table 5: Buck-Boost).
Select freewheeling diode to meet t rr determined in Step 3
(Table 3).
For direct feedback designs, if the minimum load <3 mA then
calculate R PL = V O / 3 mA.
Select C O as 100 μF , 1.25 × V O , low ESR type. Construct prototype and verify design.
2.3.4.5.6.7.
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Table 2. LinkSwitch-TN Circuit Con? gurations Using Optocoupler Feedback.
Part Number
V RRM I F t rr Package
Manufacturer
(V)
(A)
(ns)
MUR160600150Leaded Vishay UF4005600175Leaded Vishay BYV26C 600130Leaded Vishay/Philips
FE1A 600135Leaded Vishay STTA10 6600120Leaded ST Microelectronics STTA10 6U 600120SMD ST Microelectronics
US1J
600
1
75
SMD
Vishay
Table 3. List of Ultra-Fast Diodes Suitable for Use as the Freewheeling Diode.
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V OUT I OUT(MAX)
Inductor
LNK30X
Mode
Diode t rr
R FB *V Z
μH I RMS (mA)
Tokin
Coilcraft
5≤6580120160175225280360 1200 70 1200 80 680 220 680 230 680 320 680 340 680 440 680 430--SBC2-681-211SBC2-681-211SBC3-681-211SBC4-681-211SBC4-681-211SBC4-681-211
RFB0807-122RFB0807-122RFB0807-681RFB0807-681RFB0810-681RFB0810-681RFB0810-681RFB0810-681
LNK302MDCM CCM ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤35 ns 3.84 k Ω 3.9 V
LNK304MDCM CCM LNK305MDCM CCM LNK306MDCM CCM 12
≤608085120160175225280360 1800 70 2700 80 680 180 1000 230 1500 320 680 340 1000 440 680 430 1500 400--SBC2-681-211
SBC3-102-281
SBC3-152-251
SBC3-681-361SBC4-102-291
SBC4-681-431
SBC6-152-451
RFB0807-222RFB0807-272RFB0807-681RFB0807-102RFB0810-152 RFB0810-681
RFB0810-102RFB0810-681RFB1010-152LNK302MDCM CCM ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤35 ns 11.86 k Ω11 V
LNK304MDCM MDCM CCM LNK305MDCM CCM LNK306MDCM CCM 15≤658070120160175225280360 2200 70 3300 80 680 160 1200 210 1800 210 820 310 1200 310 820 390 1500 390SBC3-222-191
SBC3-332-151
SBC2-681-211
-----SBC6-152-451
RFB0807-222RFB0807-332RFB0807-681RFB0807-122 RFB0810-182 RFB0810-821RFB1010-122RFB1010-821RFB1010-152LNK302MDCM CCM ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤35 ns 15.29 k Ω13 V
LNK304MDCM MDCM CCM LNK305MDCM CCM LNK306MDCM CCM 24≤658050120160175225280360
3300 70 4700 80 680 130 1500 190 2200 180 1200 280 1500 280 1200 350 2200
360
SBC3-332-151SBC3-472-181
SBC2-681-211
SBC4-152-221
SBC4-222-211
-SBC6-152-451-SBC6-222-351
RFB0807-332RFB0807-472RFB0807-681RFB0810-152RFB0810-222RFB0810-122 RFB1010-152 RFB1010-122-LNK302MDCM CCM ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤35 ns
25.6 k Ω22 V
LNK304MDCM MDCM CCM LNK305MDCM CCM LNK306
MDCM CCM
Other Standard Components
R BIAS : 2 k Ω, 1%, 1/8 W C BP : 0.1 μF , 50 V Ceramic C FB : 10 μF , 1.25 × V O D FB : 1N4005GP R Z : 470 Ω to 2 k Ω, 1/8 W, 5%
Table 4. Components Quick Select for Buck Converters.
*Select nearest standard or combination of standard values.
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V OUT I OUT(MAX)
Inductor
LNK30X
Mode
Diode t rr
R FB *V Z
μH I RMS (mA)
Tokin
Coilcraft
5≤6580120160175225280360 1200 70 1500 80 680 220 680 230 680 340 680 320 680 440 680 430-SBC3-152-251SBC2-681-211SBC2-681-211SBC3-681-361SBC4-681-431SBC4-681-431SBC4-681-431RFB0807-122RFB0807-152RFB0807-681RFB0807-681RFB0810-681RFB0810-681RFB0810-681RFB0810-681LNK302MDCM CCM ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤35 ns 3.84 k Ω 3.9 V
LNK304MDCM CCM LNK305MDCM CCM LNK306MDCM CCM 12≤608085120160175225280360 2200 70 3300 90 680 180 1200 220 1800 210 820 320 1200 310 820 410 1800 410SBC3-222-191SBC3-332-151SBC2-681-211
------
RFB0807-222RFB0807-332RFB0807-681RFB1010-122RFB0807-182RFB0807-821RFB0810-122RFB0810-821RFB1010-182
LNK302MDCM CCM ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤35 ns 11.86 k Ω11 V
LNK304MDCM MDCM CCM LNK305MDCM CCM LNK306MDCM CCM 15
≤658070120160175225280360 2200 70 3900 90 680 180 1500 220 2200 220 1000 320 1500 320 1200 400 2200 410SBC3-222-191
-SBC2-681-211
SBC3-152-251
SBC4-222-211
SBC4-102-291
SBC4-152-251
-SBC6-222-351RFB0807-222RFB0807-392RFB0807-681RFB0807-152RFB0810-222RFB0810-102RFB0810-152RFB0810-122 RFB1010-222 LNK302MDCM CCM ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤35 ns 15.29 k Ω13 V
LNK304MDCM MDCM CCM LNK305MDCM CCM LNK306MDCM CCM 24≤658050120160175225280360
3300 70 6800 100 680 180 2200 210 3300 210 1800 300 2200 290 1800 370 3300
410
SBC3-332-151
SBC3-682-111SBC2-681-211SBC3-222-191SBC4-332-161
-SBC4-222-211
--RFB0807-332RFB0807-682RFB0807-681RFB0810-222RFB0810-332RFB0810-182RFB1010-222RFB1010-182
-
LNK302MDCM CCM ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤75 ns ≤35 ns
25.6 k Ω22 V
LNK304MDCM MDCM CCM LNK305MDCM CCM LNK306
MDCM CCM
Other Standard Components R BIAS : 2 k Ω, 1%, 1/8 W C BP : 0.1 μF , 50 V Ceramic C FB : 10 μF , 1.25 × V O D FB : 1N4005GP R Z : 470 Ω to 2 k Ω, 1/8 W, 5%
Table 5. Components Quick Select for Buck-Boost Converters. *Select nearest standard or combination of standard values.
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LinkSwitch-TN Circuit Design
LinkSwitch-TN Operation
The basic circuit con? guration for a buck converter using LinkSwitch-TN is shown in Figure 1(a).
To regulate the output, an ON/OFF control scheme is used as illustrated in Table 6. As the decision to switch is made on a cycle-by-cycle basis, the resultant power supply has extremely good transient response and removes the need for control loop compensation components. If no feedback is received for 50 ms, then the supply enters auto-restart (LNK304-306 only).
Table 6. LinkSwitch-TN Operation.
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To allow direct sensing of the output voltage without the need for a reference (Zener diode or reference IC), the FB pin voltage is tightly toleranced over the entire operating temperature range. For example, this allows a 12 V design with an overall output tolerance of ±10%. For higher performance, an opto-coupler can be used with a reference as shown in Table 2. Since the optocoupler just provides level shifting, it does not need to be safety rated or approved. The use of an opto-coupler also allows ? exibility in the location of the device, for example it allows a buck converter con? guration with the LinkSwitch-TN in the low-side return rail, reducing EMI as the SOURCE pins and connected components are no longer part of the switching node.Selecting the Topology
If possible, use the buck topology. The buck topology maximizes the available output power from a given LinkSwitch-TN and inductor value. Also, the voltage stress on the power switch
and freewheeling diode and the average current through the output inductor are slightly lower in the buck topology as compared to the buck-boost topology.Selecting the Operating Mode – MDCM and CCM Operation
At the start of a design, select between mostly discontinuous conduction mode (MDCM) and continuous conduction mode (CCM) as this decides the selection of the LinkSwitch-TN
device, freewheeling diode and inductor. For maximum output current select CCM, for all other cases MDCM is recommended. Over-all, select the operating mode and components to give the lowest overall solution cost. Table 7 summarizes the trade-offs between the two operating modes.
Additional differences between CCM and MDCM include better transient response for DCM and lower output ripple (for same capacitor ESR) for CCM. However these differences, at the low
Table 7. Comparison of Mostly Discontinuous Conduction (MDCM) and Continuous Conduction (CCM) Modes of Operation.
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output currents of LinkSwitch-TN applications, are normally not signi? cant. The conduction mode CCM or MDCM of a buck or buck-boost converter primarily depends on input voltage, output voltage, output current and device current limit. The input voltage, output voltage and output current are ? xed design parameters, therefore the LinkSwitch-TN (current limit) is the only design parameter that sets the conduction mode.
The phrase “mostly discontinuous” is used as with on-off control, since a few switching cycles may exhibit continuous inductor current, the majority of the switching cycles will be in the discontinuous conduction mode. A design can be made fully discontinuous but that will limit the available output current, making the design less cost effective.
Step-by-Step Design Procedure
Step 1. Determine System Requirements VAC MIN , VAC MAX ,
P O , V O , f L , η
Determine the input voltage range from Table 8.
Line Frequency, f L : 50 or 60 Hz, for half-wave recti?
cation use f L /2.
Output Voltage, V O : in Volts.
Output Power, P O : in Watts.Power supply ef? ciency, η: 0.7 for a 12 V output, 0.55 for a 5 V output if no better reference data available.
Step 2. Determine AC Input Stage
The input stage comprises fusible resistor(s), input recti? cation diodes and line ? lter network. The fusible resistor should be chosen as ? ameproof and, depending on the differential line input surge requirements, a wire-wound type may be required. The fusible resistor(s) provides fuse safety, inrush current limiting and differential mode noise attenuation.
For designs ≤1 W, it is lower cost to use half-wave recti? cation; >1 W, full wave recti? cation (smaller input capacitors). The EMI performance of half-wave recti? ed designs is improved by adding a second diode in the lower return rail. This provides EMI gating (EMI currents only ? ow when the diode is conducting) and also doubles differential surge withstand as the surge voltage is shared across two diodes. Table 9 shows the recommended input stage based on output power for a
universal input design while Table 10 shows how to adjust the input capacitance for other input voltage ranges.
Input (VAC)VAC MIN
VAC MAX 100/11585132230195265Universal
85
265
Table 8. Standard Worldwide Input Line Voltage Ranges.
Total Capacitance C IN(TOTAL) μF/P OUT (C IN1 + C IN2)
AC Input Voltage
(VAC)
Half Wave Recti? cation
Full Wave
Recti? cation
100/1156-83-42301-21Universal
6-8
3-4
Table 10. Suggested Total Input Capacitance Values for Different
Input Voltage Ranges.
Table 9. Recommended AC Input Stages For Universal Input.
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Step 3. Determine Minimum and Maximum DC Input Voltages V MIN and V MAX Based on AC Input Voltage Calculate V MAX as
V V 2MAX AXMAX #= (1)
Assuming that the value of input fusible resistor is small, the voltage drop across it can be ignored.
Assume bridge diode conduction time of t c = 3 ms if no other data available.
Derive minimum input voltage V MIN
V MIN =
If V MIN is ≤70 V then increase value of C IN(TOTAL).
Step 4. Select LinkSwitch-TN Device Based on Output Current and Current Limit
Decide on the operating mode - refer to Table 7.
For MDCM operation, the output current (I O ) should be less than or equal to half the value of the minimum current limit of the chosen device from the data sheet. I I 2>_LIMIT MIN O # (3)For CCM operation, the device should be chosen such that the output current I O , is more than 50%, but less than 80% of the minimum current limit I LIMIT_MIN . ..I I I 0508<<__LIMIT MIN O LIMIT MIN ## (4)Please see the data sheet for LinkSwitch-TN current limit values.
Step 5. Select the Output Inductor
Tables 4 and 5 provide inductor values and RMS current ratings for common output voltages and currents based on the
calculations in the design spreadsheet. Select the next nearest higher voltage and/or current above the required output speci? cation. Alternatively, the PIXls spreadsheet tool in the PI Expert software design suite or Appendix A can be used to calculate the exact inductor value (Eq. A7) and RMS current rating (Eq. A21).
It is recommended that the value of inductor chosen should be closer to L TYP rather than 1.5 × L TYP due to lower DC resistance and higher RMS rating. The lower limit of 680 μH limits the maximum di/dt to prevent very high peak current values. Tables 3 and 4 provide reference part numbers for standard inductors from two suppliers. 680 1.5H L L L << dependent on the inductor value. The mode of operation is a function of load current and current limit of the chosen device. The inductor value merely sets the average switching frequency. Figure 2 shows a typical standard inductor manufacturer’s data sheet. The value of off-the-shelf “drum core / dog bone / I core” inductors will drop up to 20% in value as the current increases. The constant K L_TOL in equation (A7) and the design spreadsheet adjusts for both this drop and the initial inductance value tolerance. For example if a 680 μH, 360 mA inductor is required, referring to Figure 2, the tolerance is 10% and an estimated 9.5% for the reduction in inductance at the operating current (approximately [0.36/0.38] × 10). Therefore the value of K L_TOL = 1.195 (19.5%).If no data is available, assume a K L_TOL of 1.15 (15%). PI-3783-121404 Figure 2. Example of Standard Inductor Data Sheet. Rev. F 04/09 https://www.wendangku.net/doc/1f295987.html, AN-37 Not all the energy stored in the inductor is delivered to the load, due to losses in the inductor itself. To compensate for this, a loss factor K LOSS is used. This has a recommended value of between 50% and 66% of the total supply losses as given by Equation 6. For example, a design with an overall ef? ciency (η) of 0.75 would have a K LOSS value of between 0.875 and 0.833. K to 12113 21LOSS h h =----b ^c l h m (6)Step 6. Select Freewheeling Diode For MDCM operation at t AMB ≤70 °C, select an ultra-fast diode with t rr ≤75 ns. At t AMB >70 °C, t rr ≤ 35 ns. For CCM operation, select an ultra-fast diode with t rr ≤35 ns.Allowing 25% design margin for the freewheeling diode, PIV .V V 125>MAX # (7)The diode must be able to conduct the full load current. Thus .I I 125>F O # (8)Table 3 lists common freewheeling diode choices.Step 7. Select Output Capacitor The output capacitor should be chosen based on the output voltage ripple requirement. Typically the output voltage ripple is dominated by the capacitor ESR and can be estimated as: ESR I V MAX LIMIT RIPPLE = (9) where V RIPPLE is the maximum output ripple speci? cation and I LIMIT is the LinkSwitch-TN current limit. The capacitor ESR value should be speci? ed approximately at the switching frequency of 66 kHz. Capacitor values above 100 μF are not recommended as they can prevent the output voltage from reaching regulation during the 50 ms period prior to auto-restart. If more capacitance is required, then a soft-start capacitor should be added (see Other Information section). Step 8. Select the Feedback Resistors The values of R FB and R BIAS are selected such that, at the regulated output voltage, the voltage on the FEEDBACK pin (V FB ) is 1.65 V. This voltage is speci? ed for a FEEDBACK pin current (I FB ) of 49 μA. Let the value of R BIAS = 2 k Ω; this biases the feedback network at a current of ～0.8 mA. Hence the value of R FB is given by 1.7481.652V V k R R V I V V V I R V V V R FB BIAS FB FB O FB FB FB BIAS O FB O BIAS ###X =+-=+-= -]]]g g g (10) Step 9. Select the Feedback Diode and Capacitor For the feedback capacitor, use a 10 μF general purpose electrolytic capacitor with a voltage rating ≥1.25 × V O . For the feedback diode, use a glass passivated 1N4005GP or 1N4937GP device with a voltage rating of ≥1.25 × V MAX .Step 10. Select Bypass Capacitor Use 0.1 μF, 50 V ceramic capacitor.Step 11. Select Pre-load Resistor For direct feedback designs, if the minimum load <3 mA, then calculate R PL = V O / 3 mA. Other information Startup Into Non-Resistive Loads If the total system capacitance is >100 μF or the output voltage is >12 V, then during startup the output may fail to reach regulation within 50 ms, triggering auto-restart operation. This may also be true when the load is not resistive, for example, the output is supplying a motor or fan. This is not applicable for the LNK302 as it does not have the auto-restart function. To increase the startup time, a soft-start capacitor can be added across the feedback resistor, as shown in Figure 3. The value of this soft-start capacitor is typically in the range of 0.47 μF to 47 μF with a voltage rating of 1.25 × V O . Figure 4 shows the effect of C SS used on a 12 V, 150 mA design driving a motor load. Generating Negative and Positive Outputs In appliance applications there is often a requirement to generate both an AC line referenced positive and negative output. This can be accomplished using the circuit in Figure 5. The two Zener diodes have a voltage rating close to the required output voltage for each rail and ensure that regulation is maintained when one rail is lightly and the other heavily loaded. The LinkSwitch-TN circuit is designed as if it were a single output voltage with an output current equal to the sum of both outputs. The magnitude sum of the output voltages in this example being 12 V. Figure 3. Example Schematic Showing Placement of Soft-Start Capacitor. Rev. F 04/09 https://www.wendangku.net/doc/1f295987.html, AN-37 2.5 5 28610 4Time (s) 1214 0-2V o l t a g e (V ) No soft-start capacitor. Output never reaches regulation (in auto-restart). P I -3785-010504 2.5 5 286 10 4Time (s) 12140-2V o l t a g e (V ) Soft-start capacitor value too small – output still fails to reach regulation before auto-restart. P I -3786-010504 Correct value of soft-start capacitor – output reaches regulation before auto-restart. 2.5 5 28610 4Time (s) 12140-2V o l t a g e ( V ) P I -3787-010503 Figure 4. Example of Using a Soft-Start Capacitor to Enable Driving a 12 V , 0.15 A Motor Load. All Measurements Were Made at 85 VAC (Worst Case Condition). Figure 6. High-Side Buck-Boost Constant Current Output Con? guration. Figure 5. Example Circuit – Generating Dual Output Voltages. Constant Current Circuit Con? guration (LED Driver) The circuit shown in Figure 6 is ideal for driving constant current loads such as LEDs. It uses the tight tolerance and temperature stable FEEDBACK pin of LinkSwitch-TN as the reference to provide an accurate output current. To generate a constant current output, the average output current is converted to a voltage by resistor R SENSE and capacitor C SENSE and fed into the FEEDBACK pin via R FB and R BIAS . With the values of R BIAS and R FB as shown, the value of R SENSE should be chosen to generate a voltage drop of 2 V at the required output current. Capacitor C SENSE ? lters the voltage across R SENSE , which is modulated by inductor ripple current. The value of C SENSE should be large enough to minimize the ripple voltage, especially in MDCM designs. A value of C SENSE is selected such that the time constant (t) of R SENSE and C SENSE is greater than 20 times that of the switching period (15 μs). The peak voltage seen by C SENSE is equal to R SENSE × I LIMIT(MAX).The output capacitor is optional; however with no output capacitor the load will see the full peak current (I LIMIT ) of the selected LinkSwitch-TN. Increase the value of C O (typically in the range of 100 nF to 10 μF) to reduce the peak current to an acceptable level for the load. Rev. F 04/09 https://www.wendangku.net/doc/1f295987.html, AN-37 Table 7. Inductor Voltage and Inductor Current of a Buck Converter in DCM. If the load is disconnected, feedback is lost and the large output voltage which results may cause circuit failure. To prevent this, a second voltage control loop, D FB and VR FB , can be added as shown if Figure 6. This also requires that C O is ? tted. The voltage of the Zener is selected as the next standard value above the maximum voltage across the LED string when it is in constant current operation. The same design equations / design spreadsheet can be used as for a standard buck-boost design, with the following additional considerations. V O = LED V F × Number of LEDs per string I O = LED I F × Number of strings Lower ef? ciency estimate due to R SENSE losses (enter R SENSE into design spreadsheet as inductor resistance)Set R BIAS = 2 k Ω and R FB = 300 ΩR SENSE = 2/I O C SENSE = 20 × (15 μs/R SENSE ) Select C O based on acceptable output ripple current through the load If the load can be disconnected or for additional fault protection, add voltage feedback components D FB and VR FB , in addition to C O .Thermal Environment To ensure good thermal performance, the SOURCE pin temperature should be maintained below 100 °C, by providing adequate heatsinking. For applications with high ambient temperature (>50 °C), it is recommended to build and test the power supply at the maximum operating ambient temperature and ensure that there is adequate thermal margin. The ? gures for maximum output current provided in the data sheet correspond to an ambient temperature of 50 °C and may need to be thermally derated. Also, it is recommended to use ultra-fast (≤35 ns) low reverse recovery diodes at higher operating temperatures (>70 °C).Recommended Layout Considerations Traces carrying high currents should be as short in length and thick in width as possible. These are the traces which connect the input capacitor, LinkSwitch-TN, inductor, freewheeling diode, and the output capacitor. Most off-the-shelf inductors are drum core inductors or dog-bone inductors. These inductors do not have a good closed magnetic path, and are a source of signi? cant magnetic coupling. They are a source of differential mode noise and, for this reason, they should be placed as far away as possible from the AC input lines. Appendix A Calculations for Inductor Value for Buck and Buck-Boost Topologies There is a minimum value of inductance that is required to deliver the speci? ed output power, regardless of line voltage and operating mode. 1.2.3.4.5.6.7.8.As a general case, Figure 7 shows the inductor current in discontinuous conduction mode (DCM). The following expressions are valid for both CCM as well as DCM operation. There are three unique intervals in DCM as can be seen from Figure 7. Interval t ON is when the LinkSwitch-TN is ON and the freewheeling diode is OFF. Current ramps up in the inductor from an initial value of zero. The peak current is the current limit I LIMIT of the device. Interval t OFF is when the LinkSwitch-TN is OFF and the freewheeling diode is ON. Current ramps down to zero during this interval. Interval t IDLE is when both the LinkSwitch-TN and freewheeling diode are OFF, and the inductor current is zero. In CCM, this idle state does not exist and thus t IDLE = 0.Neglecting the forward voltage drop of the freewheeling diode, we can express the current swing at the end of interval t ON in a buck converter as I t I L V V V t I I t for CCM I I t for CCM I 200>__,ON RIPPLE MIN MIN DS O ON RIPPLE LIMIT MIN O IDLE RIPPLE LIMIT MIN IDLE ##D ==--=-==]^^^g h h h (A1) where I RIPPLE = Inductor ripple current I LIMIT_MIN = Minimum current limit V MIN = Minimum DC bus voltage V DS = On state drain to source voltage drop V O = Output voltage L MIN = Minimum inductance Similarly, we can express the current swing at the end of interval t OFF as I t I L V t OFF RIPPLE MIN O OFF #D ==]g (A2) The initial current through the inductor at the beginning of each switching cycle can be expressed as I I I _INITIAL LIMIT MIN RIPPLE =- (A3) ON OFF IDLE Rev. F 04/09 https://www.wendangku.net/doc/1f295987.html, AN-37 The average current through the inductor over one switching cycle is equal to the output current I O . This current can be expressed as 1I T I I I I t t t 2121 0___O SW MAX LIMIT MIN INITIAL LIMIT MIN INITIAL ON OFF IDLE #####= ++++^^f h h p (A4) where I O = Output current. T SW_MAX = The switching interval corresponding to minimum switching frequency FS MIN . Substituting for t ON and t OFF from equations (A1) and (A2) we have I T I I I I I L V I L 12121___O SW MAX LIMIT MIN INITIAL LIMIT MIN INITIAL MIN DS O RIPPLE MIN O RIPPLE MIN ####=+++J L K K K K ^^N P O O O O h h (A5) I I I V I FS V V V V V 2_LIM LIMIT MIN INITIAL O O MIN MIN DS MIN DS O 22#####= ----^]]]h g g g (A6) For output voltages greater than 20 V, use V MAX for calculation of L MIN (Equation A6). For output voltages less than 20 V, use V MIN for calculation of L MIN to compensate for current limit delay time overshoot. This however does not account for the losses within the inductor (resistance of winding and core losses) and the freewheeling diode, which will limit the maximum power delivering capability and thus reduce the maximum output current. The minimum inductance must compensate for these losses in order to deliver speci? ed full load power. An estimate of these losses can be made by estimating the total losses in the power supply, and then allocating part of these losses to the inductor and diode. This is done by the loss factor K LOSS which increases the size of the inductor accordingly. Furthermore, typical inductors for this type of application are bobbin core or dog bone chokes. The speci? ed current rating refer to a temperature rise of 20 °C or 40 °C and to an inductance drop of 10%. We must incorporate an inductance tolerance factor K L_TOL within the expression for minimum inductance, to account for this manufacturing tolerance. The typical inductance value thus can be expressed as L I I K K V I FS V V V V V 2__TYP LIMIT MIN INITIAL L TOL LOSS O O MIN MIN DS MIN DS O 22 ######=----^]c ]h g m g (A7) where K LOSS is a loss factor, which accounts for the off-state total losses of the inductor. K L_TOL is the inductor tolerance factor and can be between 1.1 and 1.2. A typical value is 1.15. With this typical inductance we can express maximum output power as P L I I FS V V K 21___O MAX TYP LIMIT MIN INITIAL MIN MIN DS O MIN DS L TOL LOSS 22 ##### =--^h ( A8) Similarly for buck-boost topology the expressions for L TYP and P O_MAX are L I I K K V I FS 2__TYP LIMIT MIN INITIAL L TOL LOSS O O MIN 22####=-^c h m (A9) P L I I 21 __O MAX TYP LIMIT MIN INITIAL 22##=-^h (A10) Average Switching Frequency Since LinkSwitch-TN uses an on-off type of control, the frequency of switching is non-uniform due to cycle skipping. We can average this switching frequency by substituting the maximum power as the output power in Equation A8. Simplifying, we have FS L I I V I K V V V V V K 2_AVG LIMIT INITIAL O O L TOL MIN DS MIN DS O LOSS 22 #####= ----^h (A11) Similarly for buck-boost converter, simplifying Equation A9 we have FS L I I K V I K K 2_AVG LIMIT INITIAL LOSS O O LOSS L TOL 22 ####=- (A12)Calculation of RMS Currents The RMS current value through the inductor is mainly required to ensure that the inductor is appropriately sized and will not overheat. Also, RMS currents through the LinkSwitch-TN and freewheeling diode are required to estimate losses in the power supply. Assuming CCM operation, the initial current in the inductor in steady state is given by I I L V t _INITIAL LIMIT MIN O OFF #=- (A13) For DCM operation this initial current will be zero. Rev. F 04/09 https://www.wendangku.net/doc/1f295987.html, AN-37 The current through the LinkSwitch-TN as a function of time is given by 0,,0i t i t t t t I L V V V t t t SW SW ON ON INITIAL MIN DS O ON #11##==+ --]]g g (A14) The current through the freewheeling diode as a function of time is given by ,,i t t t i t I L V t t t 00_D ON D LIMIT MIN O ON SW 11##==-]]g g (A15) ,i t I L V t 00_D LIMIT MIN O #1=- ]g (A16) And the current through the inductor as a function of time is given by i t i t i t L SW D =+]]]g g g (A17)From the de? nition of RMS currents we can express the RMS currents through the switch, freewheeling diode and inductor as follows i _SW RMS =(A18) i _RMS D = (A19) i _RMS L = Since the switch and freewheeling diode currents fall to zero during the turn off and turn on intervals respectively, the RMS inductor current is simpli? ed to i i i ___D RMS L RMS SW RMS 22 =+ (A21)Table A1 lists the design equations for important parameters using the buck and buck-boost topologies. Table A1. Circuit Characteristics for Buck and Buck-Boost Topologies. Rev. F 04/09 https://www.wendangku.net/doc/1f295987.html, AN-37 Revision Notes Date A Initial Release 01/04B Corrected Tables 3 and 4.04/04C Added LNK302. 07/04D Added supplementary information to Tables 4 and 5.12/04E Corrected equation 2. 05/05F Updated Key Features column in Table 1. 04/09 For the latest updates, visit our website: https://www.wendangku.net/doc/1f295987.html, Power Integrations reserves the right to make changes to its products at any time to improve reliability or manufacturability. Power Integrations does not assume any liability arising from the use of any device or circuit described herein. POWER INTEGRATIONS MAKES NO WARRANTY HEREIN AND SPECIFICALLY DISCLAIMS ALL WARRANTIES INCLUDING, WITHOUT LIMITATION, THE IMPLIED WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE, AND NON-INFRINGEMENT OF THIRD PARTY RIGHTS.Patent Information The products and applications illustrated herein (including transformer construction and circuits external to the products) may be covered by one or more U.S. and foreign patents, or potentially by pending U.S. and foreign patent applications assigned to Power Integrations. A complete list of Power Integrations patents may be found at https://www.wendangku.net/doc/1f295987.html,. Power Integrations grants its customers a license under certain patent rights as set forth at https://www.wendangku.net/doc/1f295987.html,/ip.htm. Life Support Policy POWER INTEGRATIONS PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF POWER INTEGRATIONS. As used herein: A Life support device or system is one which, (i) is intended for surgical implant into the body, or (ii) supports or sustains life, and (iii) whose failure to perform, when properly used in accordance with instructions for use, can be reasonably expected to result in signi? cant injury or death to the user. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness.The PI logo, TOPSwitch, TinySwitch, LinkSwitch, DPA-Switch, PeakSwitch, EcoSmart, Clampless, E-Shield, Filterfuse, StakFET, PI Expert and PI FACTS are trademarks of Power Integrations, Inc. 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