bandstop filers with extended upper bassbands

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 54, NO. 6, JUNE 2006
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Bandstop Filters With Extended Upper Passbands
Ralph Levy, Life Fellow, IEEE, Richard V. Snyder, Life Fellow, IEEE, and Sanghoon Shin, Member, IEEE
Abstract—Previous microwave distributed bandstop ?lters have had their second harmonic response centered at no more than three times the fundamental bandstop mid-band frequency, due to the use of quarter-wave resonators. This limitation has now been removed by the use of compound resonators having shorter electrical length. Some con?gurations incorporate lumped capacitors, resulting in additional design degrees of freedom and wider passbands. The new theory has been developed to apply to both wide and narrow stopbands. Example ?lters having upper passband widths of up to six times the fundamental bandstop center frequency are presented. Index Terms—Bandstop ?lters, ?lters, inhomogeneous ?lters, microwave ?lters, mixed lumped distributed, noncommensurate lines, redundant lines.
Fig. 1. New commensurate bandstop ?lter. (Stubs are connected in shunt to the main lines.)
I. INTRODUCTION ANDSTOP ?lters are frequently employed to reject narrow-to-broad frequency bands located within a wide passband. Up to now, bandstop ?lters comprised of distributed elements have encountered a severe restriction on the extent of the upper passband imposed by the periodicity of the distributed elements [1]. This causes the stopbands to repeat at odd multiples of the fundamental stopband center frequency. In particular, the bandstop ?lters using quarter-wavelength resonators as previously described [1] give a ?rst upper stopband center frequency located at three times the fundamental stopband. This paper describes how this restriction may be overcome and how the ?rst upper stopband center frequency may be raised to a much higher multiple, e.g., as high as six times the fundamental stopband center frequency.1 This paper addresses the design of distributed bandstop ?lters normally realized in coaxial, stripline, or microstrip form. In [2], a capacitive-loading technique applicable to extending the width of the upper passband was proposed, with a good illustration given in [2, Figs. 1–3]. Other important references include [3], which presents a modern synthesis design technique, and [4], the results from which will be used later in this paper. Modern single-variable synthesis is used to obtain the basic rational functional form of the transfer function of a new class of commensurate-line bandstop ?lters having the desired broader upper passbands. Although exact synthesis could almost certainly be performed, a relatively new modern synthesis technique is employed, which may be termed “synthesis by
B
optimization.” This method obtains a new transfer function, using a combination of exact synthesis and approximation. This transfer function is used as a good starting point for optimization, ensuring that the optimization proceeds ef?ciently and rapidly. The optimization gives designed transfer functions having the correct number of passband ripples, for example, as predicted from the theoretical transfer function. The circuits are then modi?ed to incorporate lumped capacitors and short capacitive lines (“short” implies noncommensurate). Optimization is required also because some of the circuits employ both distributed (commensurate and noncommensurate lines) and/or lumped elements, and, presently, direct two-variable polynomial-based element extraction is not available. The combination of network synthesis and both circuit and E-M based optimization is shown to yield practical circuits having stopband widths as high as 25% of the rejection center frequency, and passbands as wide as six times the fundamental rejection-band center frequency. Example designs are given, including those capable of operating at power levels high enough to function well in dif?cult co-site scenarios (i.e., transmitters for one system located in close proximity to wideband receivers). II. THEORY A. Basic Filter Circuit The new bandstop ?lter prototype circuit is shown in Fig. 1. It consists of “compound” stubs, the th one consisting of a unit element having a relatively high impedance and a lower . These comimpedance open-circuited stub of impedance pound stub (resonant) elements are spaced by a pair of transmission lines of varying impedances and with each line being of the commensurate length. If this electrical length at resonance is 45 and the compound stubs are each of uniform impedance, , then the conventional bandstop ?lter prototype i.e., is recovered [1], [3]. The second harmonic or spurious resonance occurs when has increased to , i.e., the ratio of the center frequencies of the ?rst spurious to the fundamental ?rst harmonic is ( . When is 45 , then this ratio is 3, but when is less, the ratio increases, e.g., if is 36 , the ratio increases to 4.
Manuscript received October 24, 2005; revised February 28, 2006. R. Levy is with R. Levy Associates, La Jolla, CA 92037 USA. R. V. Snyder and S. Shin are with RS Microwave Inc., Butler, NJ 07405 USA (e-mail: r.snyder@https://www.wendangku.net/doc/2b9879902.html,). Digital Object Identi?er 10.1109/TMTT.2006.875804
1The work of Snyder and Shin [2], presented at the 2005 IMS Symposium, employed a capacitive loading technique to extend the passband, using approximate techniques rather than exact synthesis. Independent work had been carried out by the ?rst author, and it was decided to combine these efforts.
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The input impedance of the compound stub looking out from the main through line is
(1)
case. It has been found that it is necessary to include these extra unit elements in the general case in order to obtain equiripple well-matched passbands. These additional unit elements will be further discussed in Section V. The transfer matrix of the th cell is
Dividing by and expressing in terms of the Richards variusing able , obtained by substituting for (2) we then obtain where
(5)
(6) (3) The fundamental resonance condition at and are constrained by the equation maintained if is where is given by (3). Multiplication of the three matrices in (5) leads to the cell transfer matrix (7), shown at the bottom of this page. given by (3) and (6), the polynomial Substituting for form of the cell matrix becomes
(4) This condition applies to Chebyshev ?lters, but is relaxed in the case of bandstop ?lters that have an elliptic type of response where the loss poles are distributed across the stopband at various frequencies. In order for the ?lter prototype circuit to be commensurate, then, as stated previously, the connecting lines should also consist of a cascade of two unit elements of electrical length . In general, this cascaded pair no longer has a total length of 90 at resonance and, thus, does not directly approximate an ideal impedance inverter. The question then arises, can the ?lter be matched in the entire passband regions, especially that between the fundamental resonance and the ?rst harmonic? The answer is af?rmative, since the commensurate nature of the circuit means that the transfer function of the ?lter is the ratio of two rational polynomials, and, in theory, this may be synthesized to give equal ripple passband response. B. Polynomial Formation The rational polynomial form of the transfer function is derived in the following section. The ?lter is treated as a cascade of n unit cells. A typical cell (index r) consists of a compound stub connected to unit elements of impedance and on each side, as indicated in Fig. 1. Here, the subscript denotes a “main line” element. Note that the main line includes end unit and , which are not present in the convenelements , i.e., the uniform stub tional bandstop ?lter having
(8) where , and are simple polynomials in , each of degree indicated by their respective suf?ces, polynomials being even and the and with the and polynomials odd functions of The overall transfer matrix of the -cell ?lter is given by multiplication of matrices of the form given by (8), leading to the transfer matrix
(9) The and polynomials remain even, and the and polynomials odd functions of . It is seen that the real frequency loss poles are given by equating the product in the denominator to zero, i.e., the th such pole is given by
or (10) as in (4).
(7)

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The re?ection coef?cient is given by the well-known formula
(11)
(a)
zeros of distributed along the In general there are ) of the complex plane, real frequency axis (where . Note that this range corresponding to the range includes two stopbands corresponding to the loss poles occur. Hence, there are three passband regions ring at and occurring in the ranges 0 to to , in to 180. Symmetry considerations imply that there and are zeros in the range 0 to , zeros in to , and zeros in to 180. An interesting exception occurs when the ?lter is electrically symmetric, i.e., when
Symmetry condition
(12)
given by In this case, the degree of the numerator of to . The “missing” loss zero is (11) reduces from the central one in the to region, which then has zeros. Additionally, since in the symmetrical case the degree of the denominator in (11) is one greater than that , i.e., at of the numerator, one of the zeros will occur at , which is the mid-point of the the central passband to . Examples will be presented to demonstrate these characteristics. C. Synthesis by Optimization It is now fairly obvious that formal synthesis techniques should exist for these commensurate bandstop ?lter networks. However, such a development is quite a time-consuming task, especially since it will be shown that the simplest case having a Chebyshev all-pole response, i.e., with all of the loss poles coincident at one frequency in the fundamental stopband, does not give an entirely satisfactory result, and more complicated elliptic function responses are more desirable. Although formal synthesis programs have not yet been written, the certainty of their existence encouraged the development of an optimization technique using commercial optimizers.2 This has enabled various ?lter responses to be investigated, and ?lters having excellent equiripple passband responses have been derived having either Chebyshev or pseudoelliptic function stopband response. The process is facilitated by knowledge of the zero responses, as described in Section II-B. Hence the optimization method gives rapid designs having a variety of responses and is initially much simpler to develop than formal synthesis programs. An initial design may be obtained commencing from an exact prototype bandstop ?lter having uniform 90 shunt open-circuited stubs, as in [1]. Each uniform shunt stub is then replaced
2For example, “Touchstone,” which is no longer commercially available but remains in widespread use.
(b) Fig. 2. (a) Conventional prototype bandstop ?lter. (Impedance values shown,  = 45 at resonance). (b) Performance of conventional bandstop ?lter.
by a compound stub by equating the reactance slope parameters of the two circuits, which for the 90 stub is [1, Fig. 5.08-1] (13) where the stub impedance is denoted by . The reactance slope parameter for the compound stub is given by differentiation of (1), and, at the resonant angle , this leads to (14) where (4) is used to give (14) in remarkably simple form. It is . seen that (14) degenerates to (13) when The new ?lter design now proceeds by equating (13) and (14), giving (15) is, of course, given by (4). and the value of The above procedure is applied to each stub of the exact prototype ?lter. It will be illustrated here by presenting an example stubs. The basic prototype has a fractional bandwith width of 33.95% and a ripple VSWR of 1.2:1 or a return loss of 20.83 dB, and the circuit is depicted in Fig. 2(a). The performance of this bandstop ?lter with the electrical length of the at the fundamental resonance, half unit element being is shown in Fig. 2(b). It is seen that the second harmonic occurs

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(a)
(a)
(b) Fig. 3. (a) Chebyshev prototype bandstop ?lter prior to optimization.  25:7 . (b) Performance of Chebyshev ?lter prior to optimization.
(b)
=
at the expected three times the fundamental bandstop mid-band frequency, i.e., the ratio of the frequencies of the second to the ?rst harmonic is the normal 3:1. It is now desired to double this ratio to 6:1, which means that the commensurate angle must be reduced from 45 to 25.7 . Application of (15) and (4) leads to the preliminary circuit shown in Fig. 3(a) that has the response shown in Fig. 3(b). The ratio of the second to ?rst harmonic stopbands is , as desired, but the equiripple characteristics are distorted, but not to a severe degree, as the return loss remains better than 10 dB. In some cases, the performance obtained may be acceptable without further optimization, as demonstrated further in Section IV. The ideal match may be recovered almost exactly by optimization, allowing the impedances of the main lines, i.e., of Fig. 1, and the impedance of the stubs, i.e., the of the Fig. 1, to vary. In the Chebyshev case, the stub impedances are constrained by (4) so that the resonances occur at . The resulting circuit is shown in Fig. 4(a), and the response is shown zeros in Fig. 4(b). Note that there are the predicted in the main central passband. Fig. 4(c) is an expanded view of ) 0–1.6, Fig. 4(b) for the range of normalized frequency ( zeros in the and it is seen that there are the expected lower passband. The ?lter characteristic is symmetrical about ). 3.5 times normalized frequency ( A disadvantage of the new ?lter is the rather poor upper stopband skirt as depicted clearly in Fig. 4(b) and (c). This may be contrasted with the perfectly symmetrical response of the prototype ?lter shown in Fig. 2(b). Considerably improved characteristics may be obtained by allowing the loss poles to spread
(c) Fig. 4. (a) Optimized Chebyshev ?lter.  = 25:7 . (b) Performance of optimized Chebyshev ?lter. (c) Expanded view of (b), 0 < f =f o < 1:6.
out at different frequencies across the stopband, resulting in a pseudoelliptic ?lter having equiripple stopband rejection. Such a design is shown in Fig. 5(a), where the circuit was constrained to be electrically symmetric, and the rejection level was set to be 40 dB. The performance shown in Fig. 5(b) indicates that the ripple level is not absolutely ideal, which is rarely the case for passan optimization, but it is very close to this. The band re?ection zeros (or minima in this case) are retained in the range between the two stopbands. An expanded view of the lower pass and stop bands is shown in Fig. 5(c). The stopband ) of 0.94, 1, loss poles occur at normalized frequencies (

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(a)
(a)
(b)
(b)
(c) Fig. 5. (a) Pseudoelliptic symmetric bandstop ?lter.  = 25:7 . (b) Performance of pseudoelliptic symmetric bandstop ?lter.  = 25:7 . (c) Expanded view of Fig. 5(b), 0 < f =f o < 1:6.
(c) Fig. 6. (a) Full elliptic asymmetric bandstop ?lter.  = 25:7 . (b) Performance of full elliptic asymmetric bandstop ?lter.  = 25:7 . (c) Expanded view of Fig. 6(b), 0 < f =f o < 1:6.
and 1.1, in agreement with the values for the stub impedances given in Fig. 5(a), with application of (4). The lower passband shows four zeros clearly, but it is fairly obvious that the “lost” zero is either coincident with one of the four shown or would appear with further optimization. This type of slightly nonoptimal behavior is frequently demonstrated in optimization techniques, where the optimization terminates when a speci?cation is obtained closely rather than giving perfect agreement with a design theory. Fig. 5(c) in particular demonstrates the considerably improved upper stopband rejection compared with that shown in Fig. 4(c).
In another transformation of the original prototype of Fig. 2(a), a full elliptic function ?lter was designed by allowing the circuit to become electrically asymmetric so that the ?ve loss poles were allowed to separate distinctly across the stop band. The circuit is shown in Fig. 6(a), and the performance is given in Fig. 6(b) and (c). Here we see that the tenth minimum has appeared, which is in agreement with the theory of presented in Section II-B. The expanded view of Fig. 6(c) demonstrates the ?ve loss poles occurring at normalized fre) of 0.90, 0.91, 0.96, 1.04, and 1.09. There quencies (

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Fig. 7. Shunt compound stub equivalence relating: (a) a unit element cascaded with the series-connected compound stub, (b) the compound stub replaced by a series connection of open and short-circuited stubs connected in shunt with the main line, and (c) the parallel-coupled-line realization.
Fig. 8. Series compound stub equivalence relating: (a) a unit element cascaded with the shunt-connected compound stub, (b) the compound stub replaced by a parallel connection of open and short-circuited stubs connected in shunt, with the main line, and (c) the parallel-coupled-line realization.
are four distinct minima in the lower passband, and it is apparent that the ?fth one is “hidden” at about 0.4 normalized frequency. In a similar fashion, an optimization procedure is employed in the design of the wide-stopband, wide-passband ?lter (i.e., 24% stopband width, with more than 5.5:1 upper passband width), discussed as Example 2. With the initial circuit topology based on commensurate circuits (such as those illustrated in Figs. 6–10), implemented as capacitively loaded shunt resonators parallel to a stepped center conductor, the stopband upper and lower slopes are unacceptably asymmetric. Because the shunt transmission line resonators are capacitively loaded, it was hypothesized that providing similar capacitive loading on the series transmission-line portions parallel to the shunt resonators (i.e., on the through lines, clearly implementing a physically more symmetrical con?guration) would also achieve better electrical symmetry, while still achieving asymmetric loss pole placement. The resulting circuit resembles the commensurate line con?gurations derived with exact theory earlier in this paper, but includes the addition of short (i.e., lengths less than ) “redundant” capacitive sections. Several sections of the initial topology, development of the modi?ed topology, and response characteristics are illustrated in Section V. While the
argument for such a circuit modi?cation might be considered heuristic, the insight resulting in this “modi?ed topology” has proven to be effective. The modi?cation was performed with addition of short low-impedance (i.e., capacitive) transmission lines (with lengths less than 7 at fo) preceding each coupled section. These additional degrees of freedom, during optimization, allow the asymmetric placement of loss poles and the consequent achievement of essentially symmetrical attenuation slopes, both above and below the center of the rejection band. Use of the “redundant” short low-impedance lines also results in a reduction of the initially synthesized very large difference and and, thus, facilitates implementation between without very thin line sections or very small gaps. These details will be further illustrated in Section V covering example designs, but it is to be noted that the starting point for circuit modi?cation and optimization is the commensurate network resulting from exact synthesis as presented herein, with the addition of the aforementioned capacitive sections. D. Realization of Bandstop Filters of Narrow Bandwidth Using Parallel-Coupled Lines Filters having rather broad stop bandwidths may be realized directly if the impedance levels are not too high. However, in

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Fig. 9. (a) Schematic of a ?ve-section bandstop ?lter having short-circuited shunt stubs. (b) Schematic of a ?ve-section bandstop ?lter having open-circuited shunt stubs.
many cases (e.g., narrower stop bandwidths and/or small values of ), it is necessary to use resonators having loose couplings to the main through line in order to give realizable impedances. This is certainly the case with the examples presented in Section II-C. It has been found that it is convenient to use one of two types of equivalent circuits of the compound stub combined with an adjacent unit element as shown in Figs. 7 and 8. Fig. 7(a) shows the shunt stub case, which is easily converted into the circuit of Fig. 7(b). The exactly equivalent parallel-coupled-line realization of this is shown in Fig. 7(c). The equivalence between the circuits of Fig. 7(b) and (c) has been given in [4, Fig. 4]. In the present representation, the ideal :1 transformer of [4] has been absorbed into the circuit elements of Fig. 7(b). Another simpli?cation is to consider only symmetrical coupled lines, so that and . Application of the equations given in [4] then leads to the design equations given in Fig. 7. The dual circuits to those given in Fig. 7 are of equal interest and are shown in Fig. 8. This dual case starts from the series-connected version of the prototype bandstop ?lter, which is shown in Fig. 8(a). For a given bandstop ?lter speci?cation, the impedances of Fig. 7(a) are equal to the admittances in Fig. 8(a). This circuit is converted into that of Fig. 8(b) with the element values shown. The ?nal conversion into the parallel coupled line form with is shown in Fig. 8(c), with the equations being derived from the expressions given in [4, Fig. 3]. It is seen that the topological differences between the circuits of Figs. 7(c) and 8(c) are the reversals of the open and short-circuited ends of the lower coupled lines on the left and of the shunt stubs on the right. However, there are also large differences in the impedance levels. The shorted stub of Fig. 7(c) tends to have a high impedance, while the open-circuited stub of Fig. 8(c) has a low impedance. Later, it will be shown that the open-circuited stubs may be replaced by lumped capacitors. In addition to being an important modi?cation from design and construction points of view, this substitution also gives a signi?cant increase in the width of the upper passband. It enables better element values to be achieved since the band-edge angle may be increased.
(a)
(b) Fig. 10. (a) Parallel-coupled-line n = 5 Chebyshev ?lter with open-circuited shunt stubs,  = 36 , bandwidth 11.11%. (b) Filter of (a) with the stubs replaced by lumped capacitors.
Schematic diagrams of ?ve-section ?lters of the two classes are shown in Fig. 9(a) and (b). The shunt lines between the par-

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TABLE I SHUNT NETWORK PARAMETERS [SEE FIG. 9(a)]
allel-coupled-line resonators as introduced by the prototype circuit of Fig. 1 are very useful to space these from each other to reduce cross couplings, enabling thick short-circuiting walls to ground to be incorporated. These are useful also in an optimization procedure, where their impedances and lengths are useful and effective variable parameters. III. CIRCUIT PARAMETERS AS A FUNCTION OF BANDWIDTH AND Reference has been made to the effect of the bandwidth on the impedance levels within the and band-edge angle bandstop ?lter structures. This will now be illustrated by examples having two values of the bandwidth, namely, 11.11% and 33.95%, and two values of , which are 25.7 and 36 . These give ratios of the second to ?rst harmonic bandstop resonances of 6:1 and 4:1, respectively. There are also the two types of parallel-coupled-line realizations to consider, and both are demonstrated. The results are summarized in Tables I and II. The various impedance values are de?ned mainly by reference to Fig. 9, and the even- and odd-mode impedances are derived and of Figs. 7 and 8 using the formulas from
(16)
The values of the various impedances in the stub versions of the ?lters are normalized to unity, but the values of Zoe, Zoo, and Zs are normalized to 50 in order to give a more direct impression of the realizability of the coupled-line versions. The impedances of the shunt stub protototypes are given in Table I, and, in Table II, the same numerical values appear as the for the series networks, but since they are dual networks they are now admittances. Signi?cant items that arise include the following. 1) In Table I, it is seen that the values of the stub impedances are much higher for the smaller value of and become higher for the narrower bandwidth. High values of stub impedances mean a lack of support for the coupled line in air-line coaxial realizations. 2) Table II shows that the shunt open-circuited stubs have for smaller values of , making them lower impedance more dif?cult to realize directly. Section IV explains how this problem may be alleviated by replacing them with lumped capacitances to ground. 3) Table I indicates that a direct realization of the ?lters using shunt stubs is feasible for wide bandwidth and the larger . In both tables, it is seen that the Zoe and Zoo values are less realizable for broad than narrow bandwidths. 4) The prototype circuits have physical symmetry about the center of the circuit, but the parallel-coupled-line realizations are slightly asymmetric since the individual coupled line sections are themselves physically asymmetric.

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TABLE II SERIES NETWORK PARAMETERS [SEE FIG. 9(b)]
IV. REPLACEMENT OF OPEN-CIRCUITED STUBS BY LUMPED CAPACITORS As stated above in 2), it is often advantageous to replace the open-circuited stubs in the realization of Figs. 8 or 9(b) by lumped capacitors. A capacitance value is selected to give the same susceptance to ground as the open-circuited stub at the mid-band frequency of the bandstop region, i.e., at the electrical length . It is important to note that such a selection provides a good starting point for optimization, reducing the time required to optimize this mixed (both lumped and distributed variable) circuit. with As an example, we consider the case and the 11.11% bandwidth, as this is the second design given in Table II. The performance of the initial fully distributed ?lter with stubs is shown in Fig. 10(a). This is an example where optimization may not be required, as the worst return loss ripple in the main central passband is better than 15 dB and mainly better than the 20-dB return loss level of the original prototype having uniform 90 lines. The performance after replacing the stubs with lumped capacitors is shown in Fig. 10(b). The return loss has degraded to about 13 dB, but the most interesting feature is the much widened central passband, where the second harmonic has been ) of 4 to over 5. increased from the normalized frequency ( Thus, rather than having the wider central passband, if the objective were to realize a 4:1 ratio using the lumped capacitors,
then it would be possible to increase , giving more realizable element values. The result of optimization on this mixed lumped and distributed circuit is shown in Fig. 11(a), where the return loss has improved to 18 dB over the central region up to ?ve times ), and the stopband has been alnormalized frequency ( lowed to become elliptic, as shown more clearly by the expanded plot of Fig. 11(b). Inclusion of the additional main-line transmission-line elements preceding and following the coupled-line sections allows for signi?cant improvement in both upper stopband slope and passband return loss. This will be shown in Section V. V. EXAMPLE DESIGNS A. Example 1 This is an example of a ?ve-section pseudoelliptic bandstop ?lter of the symmetrical pseudoelliptic type as shown in Fig. 5(a). In terms of the electrical length , the ?rst harmonic , and the passband edges for 20-dB return was at loss were at 30.5 and 44.5 , i.e., a fractional bandwidth of 37.33%. The wide bandwidth and rather large value of meant that the impedances as shown in Fig. 12(a) were realizable directly without requiring a parallel-coupled-line realization. The ripple level in the stopband was 68 dB from to , which is a 68-dB fractional bandwidth of 16%. The edge of the second harmonic passband , giving an upper passband extending from was at

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(a)
(a)
(b) Fig. 12. (a) Pseudoelliptic ?lter of Example 1.  = 37:35 . (b) Theoretical and measured performance of the ?lter shown in (a). Solid line: theoretical data; dotted line: measured data.
(b) Fig. 11. (a) Filter of Fig. 10(b) optimized to give improved return loss and elliptic-function response. (b) Expanded view of the ?lter of Fig. 11(a), 0:6 < f =f o < 1:3.
to , i.e., a ratio of 3.045:1. This is a , where considerable improvement over the case with the passband edges for the same fractional bandwidth would be and , giving a ratio of only 2.37:1. at The ?lter design giving the impedance values is shown in Fig. 12. The ?lter was built in suspended substrate stripline using 0.01-in-thick Rogers Duroid 5880. The stopband was centered at about 5.65 GHz, and the upper passband extended to beyond 18 GHz. The speci?ed return loss up to 18 GHz was 10 dB, and the initial measured results compared with the theory are shown in Fig. 12(b). Thus, it achieved the speci?ed performance with no iterations being required. The practical return loss may be improved using a ?eld-based optimization routine. B. Example 2 Fig. 13 illustrates development of a 13-section pseudoelliptic bandstop ?lter example using 21 long coupled lines, loaded with parallel-plate lumped capacitors. Fig. 13(a) shows the
Fig. 13. Design approach for example 2. (a) Several sections of coupled line con?guration with open circuit stubs replaced by loading capacitors and no redundant transmission lines between the coupled lines, i.e., “initial topology” as per Figs. 6–10 (see text). (b) Several sections of coupled-line con?guration with open circuit stubs [as in Fig. 9(b)] replaced by loading capacitors and with the addition of redundant transmission lines (Zm Zm Zm ), i.e., “modi?ed topology” (see text). (c) Asymmetric rejection response for no redundant transmission line [from Fig. 13(a) and (d)]. Symmetric rejection response due to the redundant transmission lines [from Fig. 13(b)].

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(a) Fig. 14. 13-section parallel-coupled-line pseudoelliptic band-rejection ?lter with parallel-plate lumped capacitors for ?lter from example 2.
initial topology, which is a commensurate circuit with asymmetric loss poles, derived as in Figs. 6–10. Fig. 13(b) shows the modi?ed topology, in which the “redundant” low-impedance lines have been included in the main line of the ?lter. Fig. 13(c) shows the asymmetric response resulting from the use of Fig. 13(a), while Fig. 13(d) shows the result of incorporating the “redundant” lines and optimizing. Fig. 14 presents photographs of the implemented ?lter. The desired ?rst notch is centered at 1.08 GHz with 24% bandwidth, at the 45-dB rejection level. The second-harmonic passband is at 6 GHz, with a resulting extended passband ratio of 5.56:1. However, due to the impedance mismatch at the end of passband near 6 GHz, the effective passband ratio is 5:1. It is important to note that the coupled-line lengths were reduced slightly (during optimization) to allow for the effects over the passband of the lead inductance required to implement the physical connection for the lumped capacitors. Each coupled-line section is connected through 6 long noncommensurate transmission lines for improved matching. Fig. 15 displays the measured performance for this noncommensurate and mixed-variable example. It is interesting to note that the measured return loss performance is actually better than the “theoretical” performance. The so-called theoretical performance is based upon circuit simulation, using the tabulated impedance and length data in Table III. This shows that the theoretical model using simple TEM approximation (as in Table III) is inadequate, because it does not fully incorporate the effects of tuning screws and other tuning elements. However, full analysis of the entire structure using electromagnetic simulation is very time consuming, and the results are satisfactory. The design was accomplished with an initial synthesized structure analyzed using Genesys,3 followed by co-simulation in Ansoft HFSS,4 (electromagnetic analysis and parameter extraction), followed
3Genesys, 4HFSS,
(b) Fig. 15. (a) Measured performance of 13-section parallel-coupled-line pseudoelliptic bandstop ?lter with plate lumped capacitors,  = 21 , bandwidth 22%, passband ratio = 5 : 1. (b) Expanded view of Fig. 14(a). Solid line: measured data; dotted line: theoretical data.
by optimization at the circuit level in Ansoft Designer.5 The design was implemented as a machined, air-slab structure and is intended for rejection of the military MIDS/JTIDS passband, at high power levels (300-W peak with a 120-W average), to alleviate certain communication co-site interference issues.
VI. CONCLUSION Bandstop ?lters of narrow or moderate rejection bandwidth having upper passbands which are much wider than those previously reported are presented, enabling systems designers to specify such designs. The new design technique is based on an exact analytical response function for the ?lters which could
5Designer,
by Eagleware. by Ansoft.
by Ansoft.

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TABLE III (FOR FIGS. 13–14) LUMPED LOADING CAPACITORS, 24% BANDWIDTH, 50-
TERMINATIONS
be used as the basis for a conventional single-variable exact synthesis to obtain element values. However, an almost equivalent design procedure which may be termed “synthesis by optimization” has been used, giving essentially identical and certainly very acceptable results. Both directly coupled stub and two types of parallel-coupled-line designs are described. One of the latter may be modi?ed by replacing its shunt open-circuited stubs by lumped capacitors, further extending the upper passband range. The procedure presented herein called “synthesis by optimization” obtains satisfactory ?lter response without the need for two-variable exact synthesis. The power-handling ability of the bandstop ?lters may be limited by the impedance levels of the open-circuited stubs or the physical design of the lumped capacitors that may be used in their place in some designs. The physical design of such capacitors includes choice and thickness of dielectric, as well as location within the physical structure. An example has been presented showing the application of the new design technique to ?lters capable of handling at least 300-W peak, 120-W average power in the 960–1220-MHz frequency range. Passband return loss values of a maximum of 10 dB have been attained in practice over a 6:1 passband ratio of the second to ?rst stopband center frequencies. There is no intrinsic limitation on the passband return loss based on the initial synthesis or modi?ed topology. In fact, levels of less than 15 to 20 dB are attained over much of the passband region, and further work on both matching stub inclusion and connector transition design is ongoing, with the goal of improving this return loss.
[3] O. P. Gupta and R. J. Wenzel, “Design tables for a class of optimum microwave bandstop ?lters,” IEEE Trans. Microw. Theory Tech., vol. MTT-18, no. 7, pp. 402–404, Jul. 1970. [4] H. C. Bell, “L-resonator bandstop ?lters,” IEEE Trans. Microw. Theory Tech., vol. 44, no. 12, pp. 2669–2672, Dec. 1996. [5] J. Reed and G. J. Wheeler, “A Method of analysis of symmetrical four port networks,” IRE Trans. Microw. Theory Tech., vol. MTT-4, pp. 246–252, Oct. 1956. Ralph Levy (SM’64–F’73–LF’99) received the B.A. and M.A. degrees in physics from Cambridge University, Cambridge, U.K., in 1953 and 1957, respectively, and the Ph.D. degree in applied sciences from London University, London, U.K., in 1966. From 1953 to 1959, he was with GEC, Stanmore, U.K., where he was involved with microwave components and systems. In 1959, he joined Mullard Research Laboratories, Redhill, U.K., where he developed a widely used technique for accurate instantaneous frequency measurement using several microwave discriminators in parallel known as digital IFM. This electronic countermeasures work included the development of decade bandwidth directional couplers and broad-band matching theory. From 1964 to 1967, he was a member of the faculty of The University of Leeds, Leeds, U.K., where he carried out research in microwave network synthesis, including distributed elliptic function ?lters and exact synthesis for branch-guide and multiaperture directional couplers. In 1967, he joined Microwave Development Laboratories, Natick, MA, as Vice President of Research. He developed practical techniques for the design of broad-band mixed lumped and distributed circuits, such as tapered corrugated waveguide harmonic rejection ?lters, and the synthesis of a variety of microwave passive components. This included the development of multioctave multiplexers in SSS, requiring accurate modeling of inhomogeneous stripline circuits and discontinuities. From 1984 to 1988, he was with KW Microwave, San Diego, CA, where he was mainly involved with design implementations and improvements in ?lter-based products. From August 1988 to July 1989, he was with Remec Inc., San Diego, CA, where he continued with advances in SSS components, synthesis of ?lters with arbitrary ?nite frequency poles, and microstrip ?lters. In July 1989, he became an independent consultant and has worked with many companies on a wide variety of projects, mainly in the ?eld of passive components, especially ?lters and multiplexers. He has authored approximately 70 papers and two books, and holds 12 patents. Dr. Levy has been involved in many IEEE Microwave Theory and Techniques Society (IEEE MTT-S) activities, including past editor of the IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES (1986–1988). He was chairman of the Central New England and San Diego IEEE MTT-S chapters, and was vice-chairman of the Steering Committee for the 1994 IEEE MTT-S International Microwave Symposium (IMS). He was the recipient of the 1997 IEEE MTT-S Career Award.
REFERENCES
[1] G. L. Matthaei, L. Young, and E. M. T. Jones, Microwave Filters, Impedance-Matching Networks, and Coupling Structures. New York: McGraw-Hill, 1964, see Ch. 12. [2] R. V. Snyder and S. Shin, “Parallel coupled line notch ?lter with wide spurious-free passbands,” in IEEE MTT-S Int. Microw. Symp. Dig., 2005, Paper TU4A-3, CD ROM.

LEVY et al.: BANDSTOP FILTERS WITH EXTENDED UPPER PASSBANDS
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Richard V. Snyder (S’58–M’63–SM’80–F’97– LF’05) received the B.S. degree from Loyola-Marymount University, Lost Angeles, CA, in 1961, the M.S. degree from the University of Southern California, Los Angeles, in 1962, and the Ph.D. degree from the Polytechnic Institute of New York, Brooklyn, NY, in 1982. He is President of RS Microwave, Butler, NJ. He teaches and advises at the New Jersey Institute of Technology (NJIT), Newark. He is a Visiting Professor with The University of Leeds, Leeds, U.K. He was previously Chief Engineer for Premier Microwave. He has authored 69 papers and two book chapters. He holds 17 patents. His interests include E-M simulation, dielectric and suspended resonators, high power notch ?lters, and active ?lters. Dr. Snyder served the North Jersey Section as chairman and 14-year chair of the IEEE Microwave Theory and Techniques (MTT)-Antennas and Propagation (AP) chapter. He is currently chair of the North Jersey Electron Device Society (EDS) and Circuits and Systems (CAS) chapters. He served as general chairman for the IEEE Microwave Theory and Techniques Society (IEEE MTT-S) 2003 International Microwave Symposium (IMS2003), Philadelphia, PA. In January 2005, he began a three-year term as an elected member of the Administrative Committee (AdCom). Within the AdCom, he serves as vice-chair of the TCC, vice-chair of the IMSCC, and chair of the Standards Committee. He is a member
of the American Physical Society, the American Association for the Advancement of Science (AAAS), and the New York Academy of Science. He is a reviewer for IEEE MTT-S publications and Microwave Journal. He is active in the IEEE MTT-S Speaker’s Bureau and the three above-mentioned AdCom committees. He served seven years as chair of MTT-8 and continues in MTT-8/TPC work. He was a two-time recipient of the Region 1 Award. He was the recipient of the 2000 IEEE Millennium Medal.
Sanghoon Shin (S’98–M’02) was born in Seoul, Korea, in 1967. He received the B.S. degree from Hanyang University, Seoul, Korea, in 1993, the M.S. degree in electrical engineering from the Polytechnic University of New York, Brooklyn, in 1996, and the Ph.D. degree in electrical engineering from the New Jersey Institute of Technology (NJIT), Newark, in 2002. In 2002, he joined RS Microwave Inc., Butler, NJ, where he is currently a Research Engineer. His research interest has focused on analysis and design of RF and microwave ?lters.

with复合结构专项练习96126

with复合结构专项练习（二） 一请选择最佳答案 1）With nothing_______to burn，the fire became weak and finally died out. A.leaving B.left C.leave D.to leave 2）The girl sat there quite silent and still with her eyes_______on the wall. A.fixing B.fixed C.to be fixing D.to be fixed 3）I live in the house with its door_________to the south.（这里with结构作定语） A.facing B.faces C.faced D.being faced 4）They pretended to be working hard all night with their lights____. A.burn B.burnt C.burning D.to burn 二：用with复合结构完成下列句子 1）_____________（有很多工作要做），I couldn't go to see the doctor. 2）She sat__________（低着头）。 3）The day was bright_____.（微风吹拂） 4）_________________________，（心存梦想）he went to Hollywood. 三把下列句子中的划线部分改写成with复合结构。 1）Because our lessons were over，we went to play football. _____________________________. 2）The children came running towards us and held some flowers in their hands. _____________________________. 3）My mother is ill，so I won't be able to go on holiday. _____________________________. 4）An exam will be held tomorrow，so I couldn't go to the cinema tonight. _____________________________.

With的用法全解

With的用法全解 with结构是许多英语复合结构中最常用的一种。学好它对学好复合宾语结构、不定式复合结构、动名词复合结构和独立主格结构均能起很重要的作用。本文就此的构成、特点及用法等作一较全面阐述，以帮助同学们掌握这一重要的语法知识。 一、 with结构的构成 它是由介词with或without+复合结构构成，复合结构作介词with或without的复合宾语，复合宾语中第一部分宾语由名词或代词充当，第二部分补足语由形容词、副词、介词短语、动词不定式或分词充当，分词可以是现在分词，也可以是过去分词。With结构构成方式如下： 1. with或without-名词/代词+形容词； 2. with或without-名词/代词+副词； 3. with或without-名词/代词+介词短语； 4. with或without-名词/代词 +动词不定式； 5. with或without-名词/代词 +分词。 下面分别举例： 1、 She came into the room，with her nose red because of cold.（with+名词+形容词，作伴随状语）

2、 With the meal over ， we all went home.（with+名词+副词，作时间状语） 3、The master was walking up and down with the ruler under his arm。（with+名词+介词短语，作伴随状语。） The teacher entered the classroom with a book in his hand. 4、He lay in the dark empty house，with not a man ，woman or child to say he was kind to me.（with+名词+不定式，作伴随状语）He could not finish it without me to help him.（without+代词 +不定式，作条件状语） 5、She fell asleep with the light burning.（with+名词+现在分词，作伴随状语） Without anything left in the with结构是许多英 语复合结构中最常用的一种。学好它对学好复合宾语结构、不定式复合结构、动名词复合结构和独立主格结构均能起很重要的作用。本文就此的构成、特点及用法等作一较全面阐述，以帮助同学们掌握这一重要的语法知识。 二、with结构的用法 with是介词，其意义颇多，一时难掌握。为帮助大家理清头绪，以教材中的句子为例，进行分类，并配以简单的解释。在句子中with结构多数充当状语，表示行为方式，伴随情况、时间、原因或条件（详见上述例句）。 1.带着，牵着…… (表动作特征)。如： Run with the kite like this.

精神分裂症的病因及发病机理

精神分裂症的病因及发病机理 精神分裂症病因：尚未明，近百年来的研究结果也仅发现一些可能的致病因素。（一）生物学因素1．遗传遗传因素是精神分裂症最可能的一种素质因素。国内家系调查资料表明：精神分裂症患者亲属中的患病率比一般居民高6.2倍，血缘关系愈近，患病率也愈高。双生子研究表明：遗传信息几乎相同的单卵双生子的同病率远较遗传信息不完全相同 的双卵双生子为高，综合近年来11项研究资料：单卵双生子同病率（56.7%），是双卵双生子同病率（12.7%）的4.5倍，是一般人口患难与共病率的35-60倍。说明遗传因素在本病发生中具有重要作用，寄养子研究也证明遗传因素是本症发病的主要因素，而环境因素的重要性较小。以往的研究证明疾病并不按类型进行遗传，目前认为多基因遗传方式的可能性最大，也有人认为是常染色体单基因遗传或多源性遗传。Shields发现病情愈轻，病因愈复杂，愈属多源性遗传。高发家系的前瞻性研究与分子遗传的研究相结合，可能阐明一些问题。国内有报道用人类原癌基因Ha-ras-1为探针，对精神病患者基因组进行限止性片段长度多态性的分析，结果提示11号染色体上可能存在着精神分裂症与双相情感性精神病有关的DNA序列。2．性格特征：约40%患者的病前性格具有孤僻、冷淡、敏感、多疑、富于幻想等特征，即内向

型性格。3．其它：精神分裂症发病与年龄有一定关系，多发生于青壮年，约1/2患者于20～30岁发病。发病年龄与临床类型有关，偏执型发病较晚，有资料提示偏执型平均发病年龄为35岁，其它型为23岁。80年代国内12地区调查资料：女性总患病率（7.07%。）与时点患病率（5.91%。）明显高于男性（4.33%。与3.68%。）。Kretschmer在描述性格与精神分裂症关系时指出：61%患者为瘦长型和运动家型，12.8%为肥胖型，11.3%发育不良型。在躯体疾病或分娩之后发生精神分裂症是很常见的现象，可能是心理性生理性应激的非特异性影响。部分患者在脑外伤后或感染性疾病后发病；有报告在精神分裂症患者的脑脊液中发现病毒性物质；月经期内病情加重等躯体因素都可能是诱发因素，但在精神分裂症发病机理中的价值有待进一步证实。（二）心理社会因素1．环境因素①家庭中父母的性格，言行、举止和教育方式（如放纵、溺爱、过严）等都会影响子女的心身健康或导致个性偏离常态。②家庭成员间的关系及其精神交流的紊乱。③生活不安定、居住拥挤、职业不固定、人际关系不良、噪音干扰、环境污染等均对发病有一定作用。农村精神分裂症发病率明显低于城市。2．心理因素一般认为生活事件可发诱发精神分裂症。诸如失学、失恋、学习紧张、家庭纠纷、夫妻不和、意处事故等均对发病有一定影响，但这些事件的性质均无特殊性。因此，心理因素也仅属诱发因

with用法归纳

with用法归纳 （1）“用……”表示使用工具，手段等。例如： ①We can walk with our legs and feet. 我们用腿脚行走。 ②He writes with a pencil. 他用铅笔写。 （2）“和……在一起”，表示伴随。例如： ①Can you go to a movie with me? 你能和我一起去看电影'>电影吗？ ②He often goes to the library with Jenny. 他常和詹妮一起去图书馆。 （3）“与……”。例如： I’d like to have a talk with you. 我很想和你说句话。 （4）“关于，对于”，表示一种关系或适应范围。例如： What’s wrong with your watch? 你的手表怎么了？ （5）“带有，具有”。例如： ①He’s a tall kid with short hair. 他是个长着一头短发的高个子小孩。 ②They have no money with them. 他们没带钱。 （6）“在……方面”。例如： Kate helps me with my English. 凯特帮我学英语。 （7）“随着，与……同时”。例如： With these words, he left the room. 说完这些话，他离开了房间。 [解题过程] with结构也称为with复合结构。是由with+复合宾语组成。常在句中做状语，表示谓语动作发生的伴随情况、时间、原因、方式等。其构成有下列几种情形： 1.with+名词(或代词)+现在分词 此时，现在分词和前面的名词或代词是逻辑上的主谓关系。 例如:1)With prices going up so fast, we can't afford luxuries. 由于物价上涨很快，我们买不起高档商品。（原因状语） 2)With the crowds cheering, they drove to the palace. 在人群的欢呼声中，他们驱车来到皇宫。（伴随情况） 2.with+名词(或代词)+过去分词 此时，过去分词和前面的名词或代词是逻辑上的动宾关系。

独立主格with用法小全

独立主格篇 独立主格，首先它是一个“格”，而不是一个“句子”。在英语中任何一个句子都要有主谓结构，而在这个结构中，没有真正的主语和谓语动词，但又在逻辑上构成主谓或主表关系。独立主格结构主要用于描绘性文字中，其作用相当于一个状语从句，常用来表示时间、原因、条件、行为方式或伴随情况等。除名词/代词+名词、形容词、副词、非谓语动词及介词短语外，另有with或without短语可做独立主格，其中with可省略而without不可以。*注：独立主格结构一般放在句首，表示原因时还可放在句末；表伴随状况或补充说明时，相当于一个并列句，通常放于句末。 一、独立主格结构： 1. 名词/代词+形容词 He sat in the front row, his mouth half open. Close to the bank I saw deep pools, the water blue like the sky. 靠近岸时，我看见几汪深池塘，池水碧似蓝天。 2. 名词/代词+现在分词 Winter coming, it gets colder and colder. The rain having stopped, he went out for a walk.

The question having been settled, we wound up the meeting. 也可以The question settled, we wound up the meeting. 但含义稍有差异。前者强调了动作的先后。 We redoubled our efforts, each man working like two. 我们加倍努力，一个人干两个人的活。 3. 名词/代词+过去分词 The job finished, we went home. More time given, we should have done the job much better. *当表人体部位的词做逻辑主语时，不及物动词用现在分词，及物动词用过去分词。 He lay there, his teeth set, his hands clenched, his eyes looking straight up. 他躺在那儿，牙关紧闭，双拳紧握，两眼直视上方。 4. 名词/代词+不定式 We shall assemble at ten forty-five, the procession to start moving at precisely eleven. We divided the work, he to clean the windows and I to sweep the floor.

精神分裂症的发病原因是什么

精神分裂症的发病原因是什么 精神分裂症是一种精神病，对于我们的影响是很大的，如果不幸患上就要及时做好治疗，不然后果会很严重，无法进行正常的工作和生活，是一件很尴尬的事情。因此为了避免患上这样的疾病，我们就要做好预防，今天我们就请广州协佳的专家张可斌来介绍一下精神分裂症的发病原因。 精神分裂症是严重影响人们身体健康的一种疾病，这种疾病会让我们整体看起来不正常，会出现胡言乱语的情况，甚至还会出现幻想幻听，可见精神分裂症这种病的危害程度。 (1)精神刺激：人的心理与社会因素密切相关，个人与社会环境不相适应，就产生了精神刺激，精神刺激导致大脑功能紊乱，出现精神障碍。不管是令人愉快的良性刺激，还是使人痛苦的恶性刺激，超过一定的限度都会对人的心理造成影响。 (2)遗传因素：精神病中如精神分裂症、情感性精神障碍，家族中精神病的患病率明显高于一般普通人群，而且血缘关系愈近，发病机会愈高。此外，精神发育迟滞、癫痫性精神障碍的遗传性在发病因素中也占相当的比重。这也是精神病的病因之一。 (3)自身：在同样的环境中，承受同样的精神刺激，那些心理素质差、对精神刺激耐受力低的人易发病。通常情况下，性格内向、心胸狭窄、过分自尊的人，不与人交往、孤僻懒散的人受挫折后容易出现精神异常。 (4)躯体因素：感染、中毒、颅脑外伤、肿瘤、内分泌、代谢及营养障碍等均可导致精神障碍，。但应注意，精神障碍伴有的躯体因素，并不完全与精神症状直接相关，有些是由躯体因素直接引起的，有些则是以躯体因素只作为一种诱因而存在。 孕期感染。如果在怀孕期间，孕妇感染了某种病毒，病毒也传染给了胎儿的话，那么，胎儿出生长大后患上精神分裂症的可能性是极其的大。所以怀孕中的女性朋友要注意卫生，尽量不要接触病毒源。 上述就是关于精神分裂症的发病原因，想必大家都已经知道了吧。患上精神分裂症之后，大家也不必过于伤心，现在我国的医疗水平是足以让大家快速恢复过来的，所以说一定要保持良好的情绪。

with复合宾语的用法(20201118215048)

with+复合宾语的用法 一、with的复合结构的构成 二、所谓"with的复合结构”即是"with+复合宾语”也即"with +宾语+宾语补足语” 的结构。其中的宾语一般由名词充当（有时也可由代词充当）；而宾语补足语则是根据 具体的需要由形容词，副词、介词短语，分词短语（包括现在分词和过去分词）及不定式短语充当。下面结合例句就这一结构加以具体的说明。 三、1、with +宾语+形容词作宾补 四、①He slept well with all the windows open.（82 年高考题） 上面句子中形容词open作with的宾词all the windows的补足语， ②It' s impolite to talk with your mouth full of food. 形容词短语full of food 作宾补。Don't sleep with the window ope n in win ter 2、with+宾语+副词作宾补 with Joh n away, we have got more room. He was lying in bed with all his clothes on. ③Her baby is used to sleeping with the light on.句中的on 是副词，作宾语the light 的补足语。 ④The boy can t play with his father in.句中的副词in 作宾补。 3、with+宾语+介词短语。 we sat on the grass with our backs to the wall. his wife came dow n the stairs,with her baby in her arms. They stood with their arms round each other. With tears of joy in her eyes ,she saw her daughter married. ⑤She saw a brook with red flowers and green grass on both sides. 句中介词短语on both sides 作宾语red flowersandgreen grass 的宾补， ⑥There were rows of white houses with trees in front of them.，介词短语in front of them 作宾补。 4、with+宾词+分词（短语 这一结构中作宾补用的分词有两种，一是现在分词，二是过去分词，一般来说，当分词所表 示的动作跟其前面的宾语之间存在主动关系则用现在分词，若是被动关系，则用过去分词。 ⑦In parts of Asia you must not sit with your feet pointing at another person.（高一第十课），句中用现在分词pointing at…作宾语your feet的补足语，是因它们之间存在主动关系，或者说point 这一动作是your feet发出的。 All the after noon he worked with the door locked. She sat with her head bent. She did not an swer, with her eyes still fixed on the wall. The day was bright,with a fresh breeze（微风）blowing. I won't be able to go on holiday with my mother being ill. With win ter coming on ,it is time to buy warm clothes. He soon fell asleep with the light still bur ning. ⑧From space the earth looks like ahuge water covered globe,with a few patches of land stuk ing out above the water而在下面句子中因with的宾语跟其宾补之间存在被动关系，故用过去分词作宾补：

with用法小结

with用法小结 一、with表拥有某物 Mary married a man with a lot of money . 马莉嫁给了一个有着很多钱的男人。 I often dream of a big house with a nice garden . 我经常梦想有一个带花园的大房子。 The old man lived with a little dog on the lonely island . 这个老人和一条小狗住在荒岛上。 二、with表用某种工具或手段 I cut the apple with a sharp knife . 我用一把锋利的刀削平果。 Tom drew the picture with a pencil . 汤母用铅笔画画。 三、with表人与人之间的协同关系 make friends with sb talk with sb quarrel with sb struggle with sb fight with sb play with sb work with sb cooperate with sb I have been friends with Tom for ten years since we worked with each other, and I have never quarreled with him . 自从我们一起工作以来，我和汤姆已经是十年的朋友了，我们从没有吵过架。 四、with 表原因或理由 John was in bed with high fever . 约翰因发烧卧床。 He jumped up with joy . 他因高兴跳起来。 Father is often excited with wine . 父亲常因白酒变的兴奋。 五、with 表“带来”，或“带有,具有”，在…身上，在…身边之意

精神分裂症的病因是什么

精神分裂症的病因是什么 精神分裂症是一种精神方面的疾病，青壮年发生的概率高，一般 在16～40岁间，没有正常器官的疾病出现，为一种功能性精神病。 精神分裂症大部分的患者是由于在日常的生活和工作当中受到的压力 过大，而患者没有一个良好的疏导的方式所导致。患者在出现该情况 不仅影响本人的正常社会生活，且对家庭和社会也造成很严重的影响。 精神分裂症常见的致病因素： 1、环境因素：工作环境比如经济水平低低收入人群、无职业的人群中，精神分裂症的患病率明显高于经济水平高的职业人群的患病率。还有实际的生活环境生活中的不如意不开心也会诱发该病。 2、心理因素：生活工作中的不开心不满意，导致情绪上的失控，心里长期受到压抑没有办法和没有正确的途径去发泄，如恋爱失败， 婚姻破裂，学习、工作中不愉快都会成为本病的原因。 3、遗传因素：家族中长辈或者亲属中曾经有过这样的病人，后代会出现精神分裂症的机会比正常人要高。 4、精神影响：人的心里与社会要各个方面都有着不可缺少的联系，对社会环境不适应，自己无法融入到社会中去，自己与社会环境不相

适应，精神和心情就会受到一定的影响，大脑控制着人的精神世界， 有可能促发精神分裂症。 5、身体方面：细菌感染、出现中毒情况、大脑外伤、肿瘤、身体的代谢及营养不良等均可能导致使精神分裂症，身体受到外界环境的 影响受到一定程度的伤害，心里受到打击，无法承受伤害造成的痛苦，可能会出现精神的问题。 对于精神分裂症一定要配合治疗，接受全面正确的治疗，最好的 疗法就是中医疗法加心理疗法。早发现并及时治疗并且科学合理的治疗，不要相信迷信，要去正规的医院接受合理的治疗，接受正确的治 疗按照医生的要求对症下药，配合医生和家人，给病人创造一个良好 的治疗环境，对于该病的康复和痊愈会起到意想不到的效果。

(完整版)with的复合结构用法及练习

with复合结构 一. with复合结构的常见形式 1.“with+名词/代词+介词短语”。 The man was walking on the street, with a book under his arm. 那人在街上走着，腋下夹着一本书。 2. “with+名词/代词+形容词”。 With the weather so close and stuffy, ten to one it’ll rain presently. 天气这么闷热，十之八九要下雨。 3. “with+名词/代词+副词”。 The square looks more beautiful than even with all the light on. 所有的灯亮起来，广场看起来更美。 4. “with+名词/代词+名词”。 He left home, with his wife a hopeless soul. 他走了，妻子十分伤心。 5. “with+名词/代词+done”。此结构过去分词和宾语是被动关系，表示动作已经完成。 With this problem solved, neomycin 1 is now in regular production. 随着这个问题的解决，新霉素一号现在已经正式产生。 6. “with+名词/代词+-ing分词”。此结构强调名词是-ing分词的动作的发出者或某动作、状态正在进行。 He felt more uneasy with the whole class staring at him. 全班同学看着他，他感到更不自然了。 7. “with+宾语+to do”。此结构中，不定式和宾语是被动关系，表示尚未发生的动作。 So in the afternoon, with nothing to do, I went on a round of the bookshops. 由于下午无事可做，我就去书店转了转。 二. with复合结构的句法功能 1. with 复合结构，在句中表状态或说明背景情况，常做伴随、方式、原因、条件等状语。With machinery to do all the work, they will soon have got in the crops. 由于所有的工作都是由机器进行，他们将很快收完庄稼。（原因状语） The boy always sleeps with his head on the arm. 这个孩子总是头枕着胳膊睡觉。（伴随状语）The soldier had him stand with his back to his father. 士兵要他背对着他父亲站着。（方式状语）With spring coming on, trees turn green. 春天到了，树变绿了。（时间状语） 2. with 复合结构可以作定语 Anyone with its eyes in his head can see it’s exactly like a rope. 任何一个头上长着眼睛的人都能看出它完全像一条绳子。 【高考链接】 1. ___two exams to worry about, I have to work really hard this weekend.（04北京） A. With B. Besides C. As for D. Because of 【解析】A。“with+宾语+不定式”作状语，表示原因。 2. It was a pity that the great writer died, ______his works unfinished. (04福建) A. for B. with C. from D.of 【解析】B。“with+宾语+过去分词”在句中作状语，表示状态。 3._____production up by 60%, the company has had another excellent year. (NMET) A. As B.For C. With D.Through 【解析】C。“with+宾语+副词”在句中作状语，表示程度。

With复合结构的用法小结

With复合结构的用法小结 with结构是许多英语复合结构中最常用的一种。学好它对学好复合宾语结构、不定式复合结构、动名词复合结构和独立主格结构均能起很重要的作用。本文就此的构成、特点及用法等作一较全面阐述，以帮助同学们掌握这一重要的语法知识。 一、with结构的构成 它是由介词with或without+复合结构构成，复合结构作介词with或without的复合宾语，复合宾语中第一部分宾语由名词或代词充当，第二 部分补足语由形容词、副词、介词短语、动词不定式或分词充当，分词可以是现在分词，也可以是过去分词。With结构构成方式如下： 1. with或without-名词/代词+形容词； 2. with或without-名词/代词+副词； 3. with或without-名词/代词+介词短语； 4. with或without-名词/代词+动词不定式； 5. with或without-名词/代词+分词。 下面分别举例： 1、She came into the room，with her nose red because of cold.（with+名词+形容词，作伴随状语） 2、With the meal over ，we all went home.（with+名词+副词，作时间状语） 3、The master was walking up and down with the ruler under his arm。（with+名词+介词短语，作伴随状语。）The teacher entered the classroom with a book in his hand. 4、He lay in the dark empty house，with not a man ，woman or child to say he was kind to me.（with+名词+不定式，作伴随状语）He could not finish it without me to help him.（without+代词+不定式，作条件状语） 5、She fell asleep with the light burning.（with+名词+现在分词，作伴随状语）Without anything left in the cupboard，shewent out to get something to eat.（without+代词+过去分词，作为原因状语） 二、with结构的用法 在句子中with结构多数充当状语，表示行为方式，伴随情况、时间、原因或条件（详见上述例句）。 With结构在句中也可以作定语。例如： 1.I like eating the mooncakes with eggs. 2.From space the earth looks like a huge water-covered globe with a few patches of land sticking out above the water. 3.A little boy with two of his front teeth missing ran into the house. 三、with结构的特点 1. with结构由介词with或without+复合结构构成。复合结构中第一部分与第二部分语法上是宾语和宾语补足语关系，而在逻辑上，却具有主谓关系，也就是说，可以用第一部分作主语，第二部分作谓语，构成一个句子。例如：With him taken care of，we felt quite relieved.（欣慰）→（He was taken good care of.）She fell asleep with the light burning. →（The light was burning.）With her hair gone，there could be no use for them. →（Her hair was gone.） 2. 在with结构中，第一部分为人称代词时，则该用宾格代词。例如：He could not finish it without me to help him. 四、几点说明： 1. with结构在句子中的位置：with 结构在句中作状语，表示时间、条件、原因时一般放在

with的用法

with[wIT] prep.1.与…(在)一起，带着：Come with me. 跟我一起来吧。/ I went on holiday with my friend. 我跟我朋友一起去度假。/ Do you want to walk home with me? 你愿意和我一道走回家吗 2.(表带有或拥有)有…的，持有，随身带着：I have no money with me. 我没有带钱。/ He is a man with a hot temper. 他是一个脾气暴躁的人。/ We bought a house with a garden. 我们买了一座带花园的房子。/ China is a very large country with a long history. 中国是一个具有历史悠久的大国。3.(表方式、手段或工具)以，用：He caught the ball with his left hand. 他用左手接球。/ She wrote the letter with a pencil. 她用铅笔写那封信。4.(表材料或内容)以，用：Fill the glass with wine. 把杯子装满酒。/ The road is paved with stones. 这条路用石头铺砌。5.(表状态)在…的情况下，…地：He can read French with ease. 他能轻易地读法文。/ I finished my homework though with difficulty. 虽然有困难，我还是做完了功课。6.(表让步)尽管，虽然：With all his money, he is unhappy. 尽管他有钱，他并不快乐。/ With all his efforts, he lost the match. 虽然尽了全力，他还是输了那场比赛。7.(表条件)若是，如果：With your permission, I’ll go. 如蒙你同意我就去。8.(表原因或理由)因为，由于：He is tired with work. 他工作做累了。/ At the news we all jumped with joy. 听到这消息我们都高兴得跳了起来。9.(表时间)当…的时候，在…之后：With that remark, he left. 他说了那话就离开了。/ With daylight I hurried there to see what had happened. 天一亮我就去那儿看发生了什么事。10. (表同时或随同)与…一起，随着：The girl seemed to be growing prettier with each day. 那女孩好像长得一天比一天漂亮。11.(表伴随或附带情况)同时：I slept with the window open. 我开着窗户睡觉。/ Don’t speak with your mouth full. 不要满嘴巴食物说话。12.赞成，同意：I am with you there. 在那点上我同你意见一致。13.由…照看，交…管理，把…放在某处：I left a message for you with your secretary. 我给你留了个信儿交给你的秘书了。/ The keys are with reception. 钥匙放在接待处。14 (表连同或包含)连用，包含：The meal with wine came to ￡8 each. 那顿饭连酒每人8英镑。/ With preparation and marking a teacher works 12 hours a day. 一位老师连备课带批改作业每天工作12小时。15. (表对象或关系)对，关于，就…而言，对…来说：He is pleased with his new house. 他对他的新房子很满意。/ The teacher was very angry with him. 老师对他很生气。/ It’s the same with us students. 我们学生也是这样。16.(表对立或敌对)跟，以…为对手：The dog was fighting with the cat. 狗在同猫打架。/ He’s always arguing with his brother. 他老是跟他弟弟争论。17.(在祈使句中与副词连用)：Away with him! 带他走！/ Off with your clothes! 脱掉衣服！/ Down with your money! 交出钱来！ 【用法】1.表示方式、手段或工具等时(=以，用)，注意不要受汉语意思的影响而用错搭配，如“用英语”习惯上用in English，而不是with English。2.与某些抽象名词连用时，其作用相当于一个副词：with care=carefully 认真地/ with kindness=kindly 亲切地/ with joy=joyfully 高兴地/ with anger=angrily 生气地/ with sorrow=sorrowfully 悲伤地/ with ease=easily 容易地/ with delight=delightedly 高兴地/ with great fluency =very fluently 很流利地3.表示条件时，根据情况可与虚拟语气连用：With more money I would be able to buy it. 要是钱多一点，我就买得起了。/ With better equipment, we could have finished the job even sooner. 要是设备好些，我们完成这项工作还要快些。4.比较with 和as：两者均可表示“随着”，但前者是介词，后者是连词：He will improve as he grows older. 随着年龄的增长，他会进步的。/ People’s ideas change with the change of the times. 时代变了，人们的观念也会变化。5.介词with和to 均可表示“对”，但各自的搭配不同，注意不要受汉语意思的影响而用错，如在kind, polite, rude, good, married等形容词后通常不接介词with而接to。6.复合结构“with+宾语+宾语补足语”是一个很有用的结构，它在句中主要用作状语，表示伴随、原因、时间、条件、方式等；其中的宾语补足语可以是名词、形容词、副词、现在分词、过去分词、不定式、介词短语等：I went out with the windows open. 我外出时没有关窗户。/ He stood before his teacher with his head down. 他低着头站在老师面前。/ He was lying on the bed with all his clothes on. 他和衣躺在床上。/ He died with his daughter yet a schoolgirl. 他去世时，女儿还是个小学生。/ The old man sat there with a basket beside her. 老人坐在那儿，身边放着一个篮子。/ He fell asleep with the lamp burning. 他没熄灯就睡着了。/ He sat there with his eyes closed. 他闭目坐在那儿。/ I can’t go out with all these clothes to wash. 要洗这些衣服，我无法出去了。这类结构也常用于名词后作定语：The boy with nothing on is her son. 没穿衣服的这个男孩子是她儿子。 （摘自《英语常用词多用途词典》金盾出版社） - 1 -

精神分裂症应该怎么治疗

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