Ultra-Wideband Radar Sensors for Short-Range Vehicular Applications

Ultra-Wideband Radar Sensors for Short-Range

Vehicular Applications

Ian Gresham,Senior Member,IEEE,Alan Jenkins,Member,IEEE,Robert Egri,Member,IEEE, Channabasappa Eswarappa,Senior Member,IEEE,Noyan Kinayman,Member,IEEE,Nitin Jain,Member,IEEE, Richard Anderson,Frank Kolak,Ratana Wohlert,Shawn P.Bawell,Member,IEEE,Jacqueline Bennett,Member,IEEE,

and Jean-Pierre Lanteri,Member,IEEE

Invited Paper

Abstract—The recent approval granted by the Federal Com-munications Commission(FCC)for the use of ultra-wideband signals for vehicular radar applications has provided a gateway for the introduction of these sensors in the commercial arena as early as2004.However,the rules governing the allowable spectral occupancy create significant constraints on the sensors’opera-tion.This is further complicated by the variety of applications that these sensors are being required to fulfill.A review of the motivation for the development of these sensors is followed by a discussion of the consequent implications for waveform design and limitations on system architecture.Other practical considerations such as available semiconductor technology,packaging,and assembly techniques are reviewed,and results are presented for conventional surface-mount plastic packages illustrating their usefulness in the greater than20-GHz frequency range.Suitable antenna technology for wide-band transmission is presented that is compliant with the specific restrictions stipulated in the FCC ruling.Finally,all of these considerations are combined with the presentation of a compatible integrated-circuit-based transceiver architecture.Measured results are presented for several critical circuit components including a+12-dBm driver amplifier for the transmitter,an RF pulse generator that can produce sub-1-ns pulses at a carrier frequency of24GHz,and a single-chip homo-dyne in—phase/quadrature down-conversion receiver that has a cascaded noise figure of less than7dB.All circuit components are fabricated in SiGe.

Index Terms—Automotive,radar,SiGe,ultra-wideband(UWB).

I.I NTRODUCTION

A FTER YEARS of public hearings and rule-making de-

bates,the approval granted by the Federal Communica-tions Commission(FCC)through its modified and amended 47CFR Part15regulations allocates over7GHz of usable un-licensed spectrum between3.1–10.6GHz for communications and imaging systems based on ultra-wideband(UWB)devices [1].This ruling has been the subject of much interest and,in-deed,controversy[2]–[6].A large number of both proponents

Manuscript received August4,2003;revised May13,2004.

I.Gresham,A.Jenkins,R.Egri,C.Eswarappa,N.Kinayman,R.Anderson, R.Wohlert,S.P.Bawell,J.Bennett,and https://www.wendangku.net/doc/3f19058144.html,nteri are with Corporate Research and Development,M/A-COM,Lowell,MA01853USA.

N.Jain is with Anokiwave Inc.,San Diego,CA92130USA.

F.Kolak is with the Mitre Corporation,Bedford,MA01730USA.

Digital Object Identifier10.1109/TMTT.2004.834185and opponents of the change in the regulations have been unable to agree upon the level of maximum radiated emission levels that unlicensed users of the spectrum must adhere to in order that existing radio,communication,and other licensed services will be unaffected by harmful interference caused by UWB de-vices.However,the potential ability for UWB devices to occupy spectrum concurrently with existing systems,thereby allowing the scarce spectrum to be used more efficiently,has led to the approval being granted[1].The National Telecommunications and Information Administration(NTIA),who are responsible for the licensing and control of federal and government fre-quency bands in the U.S.,were particularly concerned with the potential cumulative effect sensors could have on the global po-sitioning system(GPS)bands,with the consequence that UWB devices are required to have very low spectral emissions in the 960–1610-MHz band[7].

In addition,although maybe less well appreciated,the ruling also approved a second allocation of an unlicensed7-GHz wide spectrum between22–29GHz that is intended exclusively for vehicular radar systems.These systems are intended to detect the location and movement of objects near a vehicle by devel-oping a continuously updated360-radar map of the vehicles’surroundings.Depending upon the sophistication of the system and its intended purpose,this would enable features that can provide,amongst other functions,collision avoidance,enhanced airbag activation,and improved road handling through interac-tion with other vehicular dynamic systems.This allocation is no less controversial than the lower frequency ruling due to its requirement for coexistence with various radio astronomy and other sensitive frequency bands.In fact,it may be more prob-lematic in that it is not just adjacent to these bands,but indeed overlaps them.As we shall see,this has had a profound influ-ence on system and waveform design,and on the perceived way that systems will be introduced,first in the U.S.,and eventually in the global marketplace.

In order to understand the rationale for such a controversial move,it is worthwhile reviewing the motivation for the intro-duction of automotive radar systems.In Section II,an overview of the requirements for and of short-range radar(SRR)sensors is used as an introduction to the details of the FCC UWB ruling for vehicular radar applications.Section III considers this ruling

0018-9480/04$20.00?2004IEEE

in the context of its implications for waveform and system de-sign,and discusses some of the tradeoffs that were considered in architecture selection.Details of sensor level operational performance and consequent system level performance are shown.Section IV highlights the technological choices—me-chanical form,semiconductors,packaging,interconnects,and interfaces—that are imposed by the architecture selection,but more importantly,by both the constraints of the marketplace and the need for the sensors to be networked with other func-tions on the vehicle.In Section V,a specific implementation of a generic integrated-circuit(IC)-based SRR is shown as an example[8]–[10]along with several examples of measured circuit-level performance to demonstrate compliance with the sensor level requirements detailed in Section III.Finally,some comments on the future of UWB radar are presented.

A.Case for Vehicular SRR

The subject of automotive radar is not new,and developments have been occurring on a regular basis,as reported in the tech-nical literature for many years[11].One of the main barriers to widespread introduction to date has been the ability to fulfill the extremely demanding technical requirements at a cost level that the average consumer will find palatable.However,signif-icant advancements have been made due to the strong support that the introduction of such sensors has from the automotive industry in general[12].They perceive that the increased safety such vehicles offer to the consumer will provide a differentiating competitive advantage.A far more important,and indeed more persuasive,argument for the introduction of smart vehicles is that any increase in safety in automobiles should have a conse-quent effect in reducing the number of fatalities that occur every year.It is,of course,this second argument that has found favor with the various governmental and regulatory agencies around the world that are required to pass the corresponding legislation. Consider,in2002,the number of occupant fatalities in road crashes in the U.S.increased for the first time since1990to an estimated figure of over42000people annually.The total number of fatalities per year in the U.S.has been relatively con-stant at around40000since1991[13],despite improvements in vehicle safety design and features.This unacceptably high level led to two major initiatives in the U.S.for the Department of Transportation to work in partnership with industry to develop and deploy effective collision avoidance products[14],[15]. In2001,the National Transportation Safety Board(NTSB) concluded that the benefits and effectiveness of collision warning systems(CWSs)and adaptive cruise control(ACC) [16]were of public importance.Independent of this,a similarly high level of fatalities in the European Union—over44000in 1998—led to the European Commission(EC)setting a goal of a50%reduction in the number of annual road deaths by2010 [17],[18].Although this ambitious target can probably not be reached through the introduction of new technology alone,the EC concluded that one of the vital steps in reaching this goal was through the introduction and implementation of Advanced Driver Assistance Systems,and that a key component of such systems was the introduction of UWB radar at24GHz[18].In September2003,the EC committed to work with the European Conference of Postal and Telecommunication

Administrations Fig.1.Collision probability as a function of increased driver reaction time for varying crash scenarios[22].

(CEPT)to remove the regulatory barriers that prevented the adoption of UWB radar in Europe to facilitate the rapid deployment of SRRs at24GHz with an eventual migration to 79GHz[19].

https://www.wendangku.net/doc/3f19058144.html,plete Environment Sensing for Automotive Vehicles Given the multiplicity of various driving and traffic sce-narios,a variety of active(that provide automatic intervention) and passive(that provide driver information)safety systems are required to give a vehicle the capability to not only perceive and understand its environment,but to also act upon it.The more important of these systems in terms of traffic and object aware-ness are largely vision—both video and infrared(IR)—and radar based[20],[21].This emphasis on introducing a variety of sensing technologies to provide a vehicle with a complete environmental awareness as a means of improving safety arises from some interesting statistics.It has been estimated that95% of all road accidents involve some human error and that,in76% of all accidents,a human is solely to blame.Associated with this is the thought-provoking statistic that almost all collisions at intersections,with oncoming traffic,and rear-end impacts, could be avoided if driver reaction time was shifted forward by2s[22],[23],as presented in Fig.1.Confirmation for this comes from a related subject of much research whereby event data recorders(i.e.,black boxes)are added to vehicles to monitor,record,display,or transmit(Tx)pre-crash,crash,and post-crash data element parameters from a vehicle[24].It is postulated that almost all of these accidents could be avoided through the introduction of CWSs and automatic vehicle interventions[18],[25]through the use of a hybrid sensor array to form a safety belt around the vehicle.

A typical schematic of one of these sensor arrays is shown in Fig.2.Radar is perceived as a key element in the sensor array due to its ability to offer an immunity to weather condi-tions that is unavailable with other technological solutions.It also offers the vehicle manufacturers a stylistic advantage over ultrasonic or video sensors in that it can be mounted behind a

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https://www.wendangku.net/doc/3f19058144.html,plete environment sensing for automobiles is achieved through a hybrid array of multiple sensor technologies and functions.The functions and typical system range of operation shown are representative only.Short-range sensing requirements for the radar sensors are envisaged to cover a variety of applications,each with differing needs for range,update rate,range resolution,and other key system parameters.

fascia that (although dissipative)can be considered transparent to the radar signal without requiring specific cutouts or sim-ilar accommodations.The longer-range ACC systems have been readily available in production vehicles for over three years,and are being increasingly supplemented by video systems to im-prove the quality of information available.In the ACC systems,a narrow beam (2or 3)is typically scanned over the front of the vehicle over an azimuth variation

of 5

to 8,and the information used to regulate its speed so that a sufficient safety margin of distance to the vehicle ahead is maintained.These systems have a necessarily narrow scanning angle (for adequate azimuth resolution of objects that may be 150m ahead),and a range resolution of 1–2m.This distance requirement also means that the high-gain antennas have a necessarily small beamwidth that inhibits their usefulness in their ability to detect and track objects at close range

(20m).Increasing the number of beams to increase the scanning angle is problematic for two reasons.Firstly,the antenna has to become more complex and,therefore,more expensive.Secondly,as the number of beams is increased,the scan rate (and,hence,dwell-time per range gate)is affected.The update rate for tracking algorithms is particularly impor-tant for very short-range operation.A further consideration is the desire to have 360coverage around the car.A single sensor is clearly not a suitable solution,and a system comprised of mul-tiple networked sensors that are distributed at appropriate loca-tions is required [26].The precise number of sensors is a func-tion of the amount of coverage required,and the corresponding coverage overlap of adjacent sensors.However,depending upon the azimuth coverage of each sensor,anywhere from 4to 16in-dividual radar sensors may be required,with placement biased

toward the front bumper,followed by the rear bumper corre-sponding to the statistically highest crash areas.Proportionally fewer sensors are required for sideways looking operation.C.Short-Range Sensors—The Requirement for UWB Radar at 24GHz

Fig.2illustrates the variety of applications that the short-range sensors are required to fulfill [27],[28].They range from simple parking aid functions to more elaborate pre-crash detec-tion,and stop-and-go or short-range cruise control functionality.The precise requirements for these applications differ between each other,and also between individual vehicle manufacturers.However,it is possible to define some typical specifications,at least for the more commonplace applications.A summary of some of the key requirements for four differing sensing func-tions is given in Table I.These represent the more immediate of the applications that automotive manufacturers want to in-troduce on vehicle platforms as a first step to the eventual in-tegrated sensor solution.Parking aids are already available on many vehicles through the use of ultrasonic sensors.However,radar sensors offer potentially superior range performance,as well as being less susceptible to inclement weather conditions,or requiring custom cutouts in the bumper fascia.The pres-ence of the radar sensor is also considered to then be a gateway to some of the more complex applications,as it can be more easily integrated into the entire sensor network and its opera-tion extended,for example,from being a simple parking aid to a blind-spot detector.One of the more demanding applications is the extension of ACC to a short-range stop-and-go function-ality.This requires not only good range resolution,but also the

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TABLE I

S UMMARY OF T YPICAL S HORT -R ANGE S ENSOR S YSTEM R EQUIREMENTS FOR

A V ARIETY OF D IFFERING A

PPLICATIONS

ability to detect and track objects such as a bicycle at ranges of over 20m initially,leading to perhaps 30–40m over time as systems become more complex and mutually dependent.The longer-range ACC systems have particular difficulty in moni-toring the corners of vehicles,for example,and are looking to the short-range sensors to provide an increased warning or re-action time for cut-in https://www.wendangku.net/doc/3f19058144.html,stly,pre-crash detection net-works are seen as being essential,but demanding,applications that are critical to improving safety.An example of such a func-tion is pre-impact sensing,whereby the closing velocity of an object that is projected to impact a vehicle may be tracked in order that the threshold voltage for airbag deployment may be dynamically adjusted.This is especially important in side-im-pact situations where extra milliseconds for airbag firing could be important.Despite the differing requirements for these appli-cations,the concept of a networked sensor system,plus the cost requirements,oblige a common sensor architecture and design that is adaptable to be used in each https://www.wendangku.net/doc/3f19058144.html,worked Sensor Operation

A distributed networked sensor structure for the radar re-sults in a system that has a hierarchical detection and tracking process.The totality of information available for compiling and updating the radar map is comprised of information from each of the discrete sensors,plus additional data from the other sen-sors that are included in the hybrid sensor network.The infor-mation available from a single sensor is,therefore,a subset of the total information required by the total radar system.Each sensor consists of an RF/IF front-end that forms the air inter-face —Tx and receive (Rx)circuitry and antennas —plus associ-ated control and signal circuits.Local digital signal processing (DSP)in each sensor is required to perform first tier detection analysis by generating a priority list of detected objects within the sensors ’field of view.The objects and their associated pa-

rameters —range,amplitude,and relative-velocity —are sent via a network to a central processor or radar decision unit (RDU).The RDU is also networked via the controller area network (CAN)-bus to accept inputs from other systems within the ve-hicle,as shown in Fig.3.The entirety of the separate sensor in-puts are then analyzed by the RDU,which performs higher level detection and tracking algorithms,and initiates appropriate de-cisions and instructions that are output to the network bus.The requirements for detection ability for each sensor are,therefore,different for the complete radar system.For example,typical

numbers for the probability of

detection

and probability of false

alarm of an object may be 0.9

and

10,respec-tively,at the individual sensor level,but are substantially im-proved at the system level.Here,we will focus on the issues of the discrete UWB SRR sensor that forms a component part of the complete networked radar.For a comprehensive discourse on the much more complex issue of how the networked system functions,see Klotz [26].

II.UWB S ENSORS AND THE FCC P ART 15R ULING The key specification common to all of the differing appli-cations that the 24-GHz sensor is required to fulfill is that of range resolution,as indicated in Table I.The accuracy of time information and,therefore,the ability to precisely determine an object ’s location,is inversely proportional to the signal band-width.The precise measurement of object movements is essen-tial for the prediction of trajectories and the prevention of false alarms.For a requirement of sub-10-cm resolution,therefore,the spectral occupancy of the main lobe of the incident wave has to be at least 3-GHz wide and more likely needs to be of the order of 4-GHz wide to allow for errors,variations,and de-sign margin.The availability of such a broad spectral frequency band has to be balanced with other practical considerations.The gain,or focusing ability,of an antenna is proportional to the antenna aperture.The antenna of each sensor,therefore,has to be small enough that an aperture of suitable electrical size can be physically mounted in the restrictive space behind a vehi-cles ’bumper fascia.In addition,as a percentile of center fre-quency,it becomes easier to produce antennas that support sig-nals of at least 4-GHz wide by increasing the frequency of op-eration.However,there are also several practical considerations that conversely argue for as low an operating frequency as pos-sible [29].The requirement for multiple sensors in a networked configuration forces the average cost per sensor to be low.De-spite the advances made in ACC radar at 77GHz,technology has not yet progressed enough such that a 77-GHz multisensor system can be cost competitive.In addition,fascia and atmo-spheric loss is far higher at 77GHz than at 24GHz —an impor-tant issue in any power-limited https://www.wendangku.net/doc/3f19058144.html,stly,it is believed by Short-range Automotive Radar frequency Allocation (SARA)1that the short-range sensor market is more likely to grow with a frequency allocation that is not dependent upon rapid technolog-ical advances.Weighing all of these factors together,the (nearly)

1SARA

is a consortium of automotive manufacturers and suppliers that was

formed in Spring 2001to coordinate frequency allocation issues for SRR sensors with the FCC,European Telecommunications Standards Institute (ETSI),and the International Telecommunication Union (ITU).

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Fig.3.Schematic representation of multiple short-range sensors connected to a local network.Each sensor passes a detected object list onto the network for analysis in the RDU.

global industrial,scientific,and medical (ISM)frequency allo-cation around 24.125GHz is ideal.

Well,almost.Unfortunately,there are several licensed frequency bands that surround the 24.125-GHz ISM band that are used for predominantly low-power and sensitive appli-cations.In particular,a global spectrum allocation between 23.6–24GHz is used for radio astronomy and remote sensing.It is this band,in particular,that has caused consternation amongst other spectrum users,particularly the European Space Agency (ESA),European Meteorological Satellite Service (EUMETSAT),Committee on Radio Astronomy Frequencies (CRAF),and others [30].Passive satellite-based sensors are used by the Earth Exploration Satellite Service (EESS)for monitoring water vapor and trace gas concentration in the Earth ’s atmosphere,and the lifting of the background noise level through the introduction of millions of albeit,low-level,radiators is seen as a cause for concern.There are also concerns about the effect of increases in wide-band spectral emissions on fixed service communications (also in the 23–26-GHz band in Europe).This has led to the specific regulations imposed by the FCC,which calls for reduced emissions at certain elevation angles above the horizon over time.The specification is written in terms of an emitted power spectral density,or effective isotropic radiated power (EIRP),which allows for either a reduction in the peak transmitted power or,as is more likely,a reduction in the elevation sidelobes of the Tx antenna.Similar proposals in Europe are at the time of writing still under consideration by the ETSI.One mitigating solution that is the subject of much research —notably by the Radarnet

consortium 2—is the proposal by SARA in conjunction with the European Radiocommunications Office (ERO)to impose a sunset date on the introduction of 24-GHz radar systems of 2014or ten years after their introduction,whichever is later.In conjunction,SRR sensors would slowly migrate in frequency to be based at 79GHz.This would set an upper limit on the maximum number of sensors produced at 24GHz and,thus,limit the increase in background noise that the EESS sensors would be exposed to.It should be noted that the corresponding spectrum at 79GHz has also yet to be allocated.

Table II summarizes the key elements of the Part 153ruling governing UWB radars for vehicular applications.The asso-ciated spectral mask is illustrated in Fig.4.To be considered an UWB device,the fractional bandwidth of the spectrum measured at the 10-dB point from the peak must be at least 20%or 500MHz,regardless of the fractional bandwidth.The vehicular radars must operate between 22–29GHz in such a way that the center frequency and the frequency at which the highest level emission occurs must be greater than 24.075GHz.Thus,to be considered UWB,the vehicular radar must have at least 500-MHz bandwidth to satisfy the regulations.Normally,the spectral density of the average emission in this band should not exceed 41.3dBm/MHz.To reduce potential interference

2Radarnet —A

European-based consortium of car manufacturers,electronics

suppliers,and researchers who are developing SRR applications for 77GHz.[Online].Available:https://www.wendangku.net/doc/3f19058144.html,.

3The Electronic Communications Committee,CEPT issued an approval no-tice for the allocation of the 77–81-GHz spectrum for Automotive Short-Range Radars on March 19,2004.

2110IEEE TRANSACTIONS ON MICROW A VE THEORY AND TECHNIQUES,VOL.52,NO.9,SEPTEMBER 2004

TABLE II

S UMMARY OF THE K EY E LEMENTS OF THE FCC R ULING G OVERNING

UWB R ADARS FOR V EHICULAR A PPLICATIONS

[1]Fig.4.Spectral mask of the UWB ruling for vehicular radar.Note that the peak radiated emission limit overlaps the 23.6–24-GHz astronomical band and,thus,has specific additional limitations imposed in this area.

with radio astronomy observations and passive earth sensing satellites,the FCC further limits the radiated emissions by requiring that,in the 23.6–24.0-GHz band,the EIRP of the antenna sidelobes beyond 30above the horizontal plane not exceed 66.3dBm/MHz until 2010,and dropping to 76.3dBm/MHz beyond 2014.In addition to the average limit of 41.3dBm/MHz,the FCC also effectively limits the peak EIRP density emission to 17dBm/MHz in a 50-MHz band around the frequency of the highest power emission.These two constraints effectively dictate a maximum duty cycle for a sensor in pulsed operation to around 0.4%to take full advantage of the average power specifications.The FCC is clearly con-cerned about the interference generated by potentially millions of radars operating simultaneously.Of course,there are also limits placed on spurious emissions;see the details in [1].One proposed solution to the restriction on elevation sidelobes is to center the spectrum of the transmitted signal above 25GHz so

that the first null of

the

spectrum falls at the restricted band,thus reducing some of the demands upon antenna design.This may well become the preferred interim solution,although early sensor production is still planned for an ISM-band center frequency.

TABLE III

S UMMARY OF THE P RINCIPAL S HORT -R ANGE S ENSOR L EVEL R EQUIREMENTS

FOR O BJECT D

ETECTION

III.S YSTEM A RCHITECTURE AND W A VEFORM D ESIGN Table III summarizes some of the more important operational and practical requirements for a generic short-range sensor.By combining an understanding of the EIRP spectral limitations with our knowledge of the operational performance require-ments of the networked radar system,it becomes possible to compare the relative merits of different sensor architectures,and select the most appropriate.

A.Sensor Architecture Considerations

One can broadly characterize radar architectures of interest in vehicular applications into three generic types,which are:1)pseudorandom noise (PN)coded continuous-wave (CW)radars;2)classical frequency-chirped systems;and 3)pulsed radars [31].Simple CW Doppler radar systems are not consid-ered here,as almost all of the vehicular applications require the detection of zero relative velocity https://www.wendangku.net/doc/3f19058144.html,mon to all of these techniques is the ability to employ a process known as pulse compression.This is a traditional technique for enhancing radar performance and may be used to improve sensitivity for detecting targets at long range.It is also possible to combine techniques to form hybrid radar systems.

Frequency chirp radars that use pulse compression are ideal for long-range higher power applications.Resolution is related to how fast and over how wide a bandwidth it is possible to generate a well-defined chirp.Close-range and closely spaced targets also put stringent requirements on local oscillator (LO)phase noise.Both of these requirements can be challenging in the UWB context in any cost-effective manner.

PN-coded radar —essentially a spread-spectrum-type system —also provides the possibility for pulse compression,often referred to as coding or processing gain in the context of PN codes.Here,flexibility exists with the choice of the code,allowing designers to trade various performance parameters,with the limitation that dynamic range is often restricted by code auto-correlation properties [28].The ability to assign different codes to systems that share the same spectrum also provides an inherent resilience to interferers making it a good

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choice for a multiradar unit system such as that envisaged for vehicular use.However,to maintain the high range resolution, a very fast set of code generators is required,with typical implementations requiring each code generator to be run at a well-defined frequency with a constant offset between one another.This would require two phased-locked loops,which could add significant cost.However,the principal disadvantage of PN-coded radars is that of dynamic range.In addition to the significant variation in the target-return signal level that occurs through the ratio of minimum to maximum range must be added the diversity of radar cross sections(RCSs)that complex targets may present to the radar system.It is extremely difficult to maintain adequate isolation between the Tx and Rx antennas such that the required dynamic range may be achieved,bearing in mind that there is a wide variation(and little influence that may be had)in the mounting environment of the sensor behind the vehicles fascia.This often restricts the operating range of the PN-coded radar to distances of less than10m.

Longer-range detection requirements force a classical pulsed radar solution to be considered.Although complex timing and delay circuitry is required,isolation between the Tx and Rx can be increased somewhat due to its time-gated nature,hence,en-hancing the dynamic range.Other desirable attributes include good range accuracy,clutter reduction,and multipath resolu-tion[31].Pulsed radar is also perhaps one of the simplest archi-tectures to implement,thus potentially making it the most cost effective.However,again the requirement of a radiated spec-trum of at least3-GHz wide proves challenging.The minimum time delay between two consecutive pulses is determined by the closest range gate.A two-way flight time of1ns(corresponding to a target at a range of0.15m)implies that either two sources are used(to generate the first IF frequency for a heterodyne re-ceiver)or a single source has to slew over at least3GHz and be stable in less than1ns.Neither is practical,which directs us toward a homodyne receiver architecture with direct conversion to baseband.The peak-power limitation of the FCC ruling also limits the amount of radiated energy that may be transmitted. This can result in short-pulse radars being limited in range.Even though the sensors are only required to detect objects at up to 20-m and eventually30-m distances,it is necessary to employ techniques such as a random pulse repetition interval jitter,in-creased dwell times for selective range gates,and pulse-com-pression techniques to maximize the amount of energy trans-mitted within the available spectrum https://www.wendangku.net/doc/3f19058144.html,stly,as a result of the zero IF structure,there is the added problem of detecting the signal in the presence of often large dc offsets.However,the in-herently high isolation between range bins that can be achieved relatively easily using switches makes this the most cost-effec-tive solution given the other challenges.

B.Time-Gated Correlation Receiver Architecture

The correlation receiver is a time-domain equivalent of the optimal frequency-domain matched filter receiver[32].Corre-lation occurs by multiplying the incident signal from the Rx an-tenna with a locally generated replica of the transmitted pulse, and integrating the result.A simplified schematic diagram of a second-generation pulsed-SRR sensor is shown in Fig.5.The RF front end is separated into three distinct sections,which are:1)Tx circuitry;2)Rx circuitry;and3)control and processing circuits.The operation of the sensor is as follows.A pulse of RF energy is transmitted from the sensor at a

time,as deter-mined by a trigger signal.This event simultaneously activates a local delay generator that commences to wait for a certain time until a second trigger is activated,which is the indication for the receiver to sample its output and store the sampled signal for further processing.Before the second trigger is activated,the Tx/Rx select switch is toggled between the Tx antenna and the LO ports of the receiver.The broad-band input signal from the Rx antenna is then multiplied with a locally generated replica of the Tx signal,resulting in pulse expansion if the two signals are time correlated.The result of the multiplication operation is in-tegrated in the baseband filter.In the pulse radar,this is done on a pulse-by-pulse basis,with the consequence that,under ideal conditions,the receiver output would be zero if the pulses do not overlap in the time domain,and a maximum that is propor-tional to the magnitude of the received pulse if they are coin-cident.As there is a direct relationship between the duration of the pulse and its spatial resolution,this coincidence in time fixes the range resolution.Thus,by varying the time delay between these two trigger events,it is possible to determine the presence of multiple objects at varying distances from the sensor corre-sponding to the minimum time delay(closest in range)to the maximum time delay(farthest in range).Of course,in reality,we have real-world issues to contend with such as wide-band noise, the presence of interferers,clutter(multiple unwanted reflec-tions),dynamic range limitations,pulse smearing,and all of the traditional issues that make receiver design challenging.Mul-tiple samples of the receiver output are necessary,therefore,to average the returned signals corresponding to a particular time delay(or range gate)and determine the fundamental question of any radar system:was an object detected or not?The prob-ability of making this decision correctly is proportional to the number of pulses collected in the receiver for each range gate. This is,in turn,a function of the gate length,which is deter-mined by the required range resolution,and update rate of the radar map.Detection algorithms that can dynamically vary the amount of dwell time per range gate and,thus,spend a propor-tionally longer time collecting signal returns from those farthest from the sensor are,therefore,beneficial.For example,a rider on a motorbike may present an RCS of only several square me-ters,as compared with a typical automobile,which may be an order of magnitude higher[33].In order to reliably detect a mo-torbike at a range of greater than20m,it may be necessary to collect and average the returns from a few hundred Tx pulses. The signal-to-noise ratio for a single pulse may be negative,and only by averaging noise over a period of time does it becomes possible to make a detection decision.

C.Interference Issues and Mitigation

A large variation in the magnitude of the returned radar signal is not the only challenge that the sensor has to contend with. There are at least three classes of interference relevant to the UW

B vehicular radar.These are:1)the spurious emissions gen-erated by the sensor and radiated as radio frequency interference (RFI);2)spurious emissions radiated in-band as a consequence of finite switch isolation;and3)in-band interferers or jammers

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2004

Fig.5.Simplified schematic diagram of the SRR RF front-end.An object at range R is detected by measuring the elapsed time between a transmitted pulse and a correlated received signal.The transceiver is split into three distinct functions.

that originate from other sensor systems.The first and only class important to regulatory agencies is the spurious emissions the radar generates.These emissions,of course,must be appropri-ately filtered in accordance with regulatory spectral emission limits.In addition,there is also a spurious emission from the pulse radar that is caused by the finite switch isolation.This is illustrated in Fig.6.Since the oscillator is continuously running,finite switch isolation results in a CW tone being continually transmitted.Although the switches used to generate the short Tx-pulse reduce the amplitude of this tone to a level that satisfies the FCC requirement on CW leakage,the signal can still cause a significant problem.The signal is reflected from the target under observation and interferes with the receiver ’s detection process by reducing its effective radar cross-sectional dynamic range;it also increases the false alarm rate and/or reduces detection probability.Similarly,the last important class of interferers to the UWB radar receiver is jamming arising from CW signals that appear in the receiving band —e.g.,from other UWB

radars

Fig.6.Measured output spectrum of the pulse former being driven to produce a 1-ns pulse with a PRF of 30MHz.The 24.15-GHz LO CW leakage can be clearly seen.

employing FM-CW-type architectures.A particularly difficult scenario is if another vehicular radar is facing the SRR only a

GRESHAM et al.:UWB RADAR SENSORS FOR SHORT-RANGE VEHICULAR APPLICATIONS 2113

few meters away.This jamming can potentially saturate the re-ceiver unless the latter is carefully designed.These interferences can be mitigated to some extent in a coherent radar by changing the carrier frequency from pulse to pulse.The radar is also sus-ceptible to other UWB pulse radars,and it can jam its own sisters mounted on the same bumper,although the time-gated opera-tion helps to an extent,as does randomizing,or staggering,the pulse epochs.A further benefit of this randomized pulsing is in meeting the spectral emission limits of the FCC by spreading the radiated energy more evenly across the operating bandwidth.

IV .S UITABLE T ECHNOLOGY FOR UWB S ENSORS

AND I NTEGRATED S ENSOR D ESIGN

Currently,sensors scheduled for production are fabricated using conventional surface-mount components and distributed transmission-line-based circuits with discrete baseband pro-cessing and control circuitry.Aside from cost,repeatability,and yield issues in following this approach,the wide instanta-neous bandwidth of sub-1-nS-long pulses makes the design of distributed circuits problematic.It is also difficult to increase or vary the functionality without increasing cost.Fig.7shows a photograph of these first-generation sensors and an exploded view of their construction.It should be noted that Fig.7does not represent an implementation of the sensor depicted in Fig.5.The historical path in the electronics industry for reducing cost,and improving and increasing functionality,has been to migrate toward IC-based solutions.Fig.5represents an evolution of the first-generation UWB sensor due to be launched in production in 2005[8].The baseband and microwave circuitry are realized using three separate ICs —a single-chip Tx IC,a single-chip Rx IC,and an application-specific integrated circuit (ASIC)that produces control signals and performs pre-processing of received data for formatting and interfacing with the vehi-cles ’CAN-bus.Separation of the Tx and Rx functions into separate ICs is a pragmatic decision based upon the operating environment.The large dynamic range required of the sensor has receiver sensitivity as one of its principal limitations.By mounting the noise sensitive LNA and receiver in a different location than the signal source and potentially noisy control signals gives the designer the most control over isolation between the two functions.The following sections provide an overview of the operation of the transceiver,followed by a brief review of the practical implications for realizing the critical circuit functions in an IC.A.Tx IC Operation

The Tx IC for the UWB sensor has a CW source that gen-erates a signal around 24.150GHz.The oscillator is required to operate in a CW mode as the time intervals between pulses do not allow for the turn-on and stabilization time of even a

low-resonator-based oscillator.Two consequences of this are the problems of LO breakthrough previously discussed,and ad-ditional self-mixing in the direct down-conversion receiver that manifests itself as a dc offset at the receiver output.It is im-portant,therefore,to maximize the isolation between the CW source and the correlator in the receiver.The voltage-controlled oscillator (VCO)also has a dc-coupled FM port that may

be

Fig.7.Exploded view of first-generation short-range sensor assembly.

used to provide frequency correction or frequency modulation.The transmitted pulse is formed from the CW source by gating the output of the oscillator to the Tx antenna for a length of time equivalent to the Tx pulse length.After the pulse has been formed,the oscillator is then directed into an absorptive load to reduce the amount of CW breakthrough.This gating opera-tion is performed by a fast switching single-pole-single-throw (SPST)absorptive switch.Varying the length of the gating con-trol signal can vary the length of the transmitted pulse.For longer pulses,this opens up the possibility of superimposing a phase-code onto the RF pulse.The principal advantage of coding in this way is that the length of the transmitted pulse (and,therefore,the incident energy upon the target)can be increased to improve the detection capability of the sensor.If the length of each bit of the code is chosen appropriately,the longer pulse can be transmitted without violating spectral restrictions,and with the same range resolution as for a very short pulse.Again,the use of pulse compression enhances radar performance in im-proving sensitivity for detecting targets at long range.There is a negative impact in that the dynamic range around each range gate is reduced as a consequence of the code sidelobes [31],but by varying the code length as a function of the range gate under consideration,this effect can be minimized [9].For very short-range applications,say,less than 5m,a simple uncoded pulse may be transmitted.Barker codes are optimal in respect of their peak-to-sidelobe ratio for any set of truncated coding sequences and,being bipolar,are easily implemented through the addition of a binary phase-shift keyed (BPSK)modulator in the Tx path.There is an important synchronization requirement between the pulse length and code generator and,in addition,to minimize the effect that distortion has upon corrupting the

2114IEEE TRANSACTIONS ON MICROW A VE THEORY AND TECHNIQUES,VOL.52,NO.9,SEPTEMBER 2004

peak-sidelobe ratio,high-speed digital circuits that switch be-tween the two phase-shift states in as short a time as possible are required.The BPSK modulator in this design is required to change phase states in approximately 50ps or a fraction of 1/10th of the bit rate.This is possible through the careful ap-plication of employing high-speed switching techniques [34].Following the BPSK modulator,a Tx/Rx select switch is used to direct the generated pulse either to the Tx antenna,or to the Rx IC where it is used as the LO drive when the sensor is in Rx https://www.wendangku.net/doc/3f19058144.html,stly,the antenna driver amplifier is used to amplify the relatively low-power signal to the maximum power level allow-able under the regulations.By optimizing the duty cycle of the transmitted pulse waveform to meet the requirements of average radiated power spectral density,this demands a peak CW power output from the driver amplifier of

approximately 12dBm.This assumes a peak antenna gain of 14dBi and includes com-pensation for packaging and interconnects loss between the am-plifier output and the antenna.A variable gain control for the driver amplifier further helps to improve the dynamic range of the sensor (again,in Rx mode)by increasing the isolation be-tween the Tx and Rx antennas.Rise and fall times of around 10ns for the amplifier to be enabled means that this facility can only be used for longer range detection.

B.Rx Chain Operation

In-phase (I)and quadrature (Q)channels are used in the re-ceiver to facilitate the extraction of Doppler information from the received signal.Both I and Q correlators (multiplier and in-tegrator)are preceded by a low-noise amplifier (LNA).It is im-perative that the first stage of the receiver is designed to have as low an added noise contribution as possible.Including pack-aging and interconnect loss,the complete cascaded noise figure of the receiver is required to be approximately 8dB in order to meet the requirements for probability of detection and max-imum false-alarm rates for the minimum RCS target at max-imum range.The I and Q signals within the receiver are created by driving the LO ports of each multiplier with an LO signal that has a relative phase shift of 90.In order that the receiver does not continually integrate broad-band noise from the Rx an-tenna and the LO signal leakage during the inter-pulse period,it is important that the output of each correlator is passed via a single-pulse sampler to the next stage of baseband filtering.Levels of noise in the receiver may be orders of magnitude higher than the voltage level of the received pulse,and coherent collection of several hundred pulses may be necessary before a decision threshold is reached.This requires two things:pre-cise timing of the sampler for the integrate and dump circuitry so that the correlated signal is sampled at its peak output,and a dc-coupled receiver.As with all direct down-conversion re-ceivers,significant signal information exists at dc and low fre-quency,although any dc offset originating from noise in the re-ceiver and LO self-mixing can mask this data and make subse-quent processing problematic.It is,therefore,necessary to oc-casionally calibrate the level of this natural dc level,and correct for it through dc regeneration in the subsequent stages of the re-ceiver.

The performance requirements of both the individual sensor and the overall sensor network are well understood.In order to meet the goal of reducing the cost of the sensors by migrating to a low-cost packaging and IC technology,some of the prac-tical considerations in manufacturing millimeter-wave sensors using low-cost techniques are the issues of the choice of a suit-able semiconductor technology:the device packaging and as-sembly techniques;and the choice of a low-loss circuit board.The key requirements to be demonstrated are that the critical electrical functions can be demonstrated in the chosen semicon-ductor technology;that conventional low-cost plastic package techniques are appropriate for high-frequency circuitry;and that conventional circuit boards and SMT procedures are compatible with the technological requirements of a planar antenna with high gain and low elevation sidelobes.

C.Semiconductor Technology —The Argument for SiGe The capability of adding SiGe epitaxial layers to a standard CMOS or BiCMOS process,so that high-speed HBTs may be integrated with conventional Si circuits,has revolutionized the course of microwave circuit design over the last few years.The ability to have denser functionality,and better control over system partitioning between the RF and digital domains,coupled with the economies of scale and portability that conventional Si fabrication offers makes extending the design of both analog and mixed-signal ICs to microwave and millimeter-wave frequencies the logical next step [35].SiGe can,therefore,offer the opportunity to provide very low-cost microwave and millimeter ICs,along with the potential to integrate digital and other control circuits adjacent to the mi-crowave and analog functions.This offers both the system and IC designer greater flexibility in system partitioning.However,SiGe is not a technological panacea,and has a number of performance limitations where it is inferior to traditional III –V semiconductors.In particular,its output power capability and noise figure —two critical transceiver design parameters —are inferior to many of the GaAs and InP devices currently available.However,the

high of many standard commercial processes proves to be adequate to compensate for these shortfalls,and critical circuit functions can be realized using SiGe.For example,in order to meet the required short-range sensor performance shown in Table III,the output power of the antenna driver amplifier only needs to be of the order from

approximately 10

to 12dBm.This is a consequence of the relatively low peak-power level stipulated by the regulatory agencies.In addition,although a lower noise-figure receiver is always desirable,the key figure-of-merit to be considered is whether or not the noise figure is simply sufficient.From Table III,a cascaded Rx noise figure of 8dB is required for a sensor to detect an object with a RCS of 10m at ranges of approximately 25m with a 1-ns https://www.wendangku.net/doc/3f19058144.html,stly,the generation of 24-GHz RF pulses of 1-ns and sub-1-ns duration is critical.All of these vital performance metrics —high-speed pulse gen-eration,transmitted signal power,and low-noise receivers —are demonstrated using a commercial low-cost SiGe foundry process and are presented in Section V .

D.Plastic Packaging for ICs at 24GHz —Addressing Cost and Manufacturing Issues

Of course,ICs that meet both cost and performance expec-tations also require complementary advances in how they are packaged and assembled,which must be both reliable and cost effective.Analogous to the desire to benefit from established economies of scale at the semiconductor level is the ability to package these semiconductors in plastic packages due to their availability and price.This is a key element in the ability to produce truly low-cost millimeter-wave systems.One pack-aging solution that offers these features are molded lead-frame (MLF)packages.They are of particular interest as they have an open lead frame as the reverse side of the package,thus offering a good,local,RF,and thermal ground.They are also compatible with automated assembly techniques.MLFs have minimal height and short vertical transitions,are robust,and are only slightly larger than the dies they protect.The size and form factor not only support high-frequency plastic package design,they lend themselves to tape and reel surface mount (SMT)assembly.In addition,packages are already available Pb-free from many manufacturers to be compliant with new environmental initiatives that come into force after July 1,2006[36].While being state-of-the-art,they are also mature enough to be considered for high-volume use.A recent advance in molding compounds with low dissipation factors is also a welcome attribute.

Until now,the use of plastic packages has been limited to relatively low frequencies (approximately 5GHz).The factors restricting the use of these packages at higher frequencies has been in part due to the lossy dielectric encapsulate,but more so to the high parasitic inductance of the bond-wire interconnects that results in high insertion loss,poor impedance matching,and a variable response over the required frequency band.Occasion-ally,this problem has been overcome by including matching stubs on the monolithic microwave integrated circuit (MMIC)chip itself [37].However,as semiconductor area is at a pre-mium for yield and cost reasons,this on-chip matching can be relatively expensive.It is also possible in some applications to use the interconnect parasitics themselves for matching the de-vices on the IC,but this is not always easy,and not recom-mended for reasons of process variation.Some package man-ufacturers have proposed flip-chip versions of MLF packages to alleviate this issue,where a bumped die is flipped onto a lead frame paddle and then molded using the standard assembly.The flip-chip orientation reduces signal inductance since the inter-connect length is much shorter than when compared to a bond wire (0.1versus 1mm).Studies have shown that this approach may provide a good broad-band transition to 40GHz [38].How-ever,the manufacturing processes for such flip-chip packages are not perfected yet,and are a way from reliable and repeat-able mass production.For narrower band applications —20%in the case of the UWB SRR sensor —it is still possible to use a standard wire-bonded version of a plastic MLF package,with compensation circuitry only included on the motherboard to compensate for the bond-wire inductance.Fig.8shows a VCO

MMIC mounted in a 20-lead 4

mm

4mm MLF package.The

18-m-diameter bond wire for the output can be

clearly

Fig.8.Measured output power of a 20–25-GHz VCO mounted in an MLF SMT package with a conventional wire-bond https://www.wendangku.net/doc/3f19058144.html,pensation for the wire-bond inductance is included on the PCB.

seen in the X-ray image.Careful package and board intercon-nect design techniques maintain a flat output power character-istic

(0.5dB including device variations)over the 20–25-GHz frequency range.

Of course,such techniques assume that the repeatability of the assembly —at both die and board level —are consistent enough to allow the variations to be absorbed into the design.As a result,M/A-COM,Lowell,MA,is working with preferred MLF packaging houses to generate die placement,die attach,lead-frame,and wire-bonding processes with the control and repeatability needed to ensure 24-GHz operation.

V .E XAMPLES OF S ENSOR C OMPONENT D ESIGN

Results are presented here that demonstrate that the key system parameters required to meet the sensor level perfor-mance metrics may be achieved using a commercial SiGe process.Other key components in SiGe at 24GHz —VCOs,switches,pre-scalars,high-speed digital circuits —are compre-hensively covered elsewhere,e.g.,[39]–[41].The ability to use MLF plastic packages for microwave ICs at frequencies above 20GHz has been shown with results presented in Section IV for a standard wire-bond transition that demonstrates good performance over the 20–25-GHz band.Also,the challenge of making a high-efficiency antenna on a representative low-cost printed circuit board (PCB)material,which meets both the bandwidth and regulatory requirements,is discussed and results presented.

An excellent review of SiGe HBT technology can be found in [42].For this work,the SiGe2rf technology of Atmel GmbH,Heilbronn,Germany,was used [43].The process is based on a

0.5-m emitter SiGe HBT with a transit frequency

of 85GHz [43].The process includes two types of HBTs on a single wafer through the use of selectively implanted collector (SIC)regions that trade a higher speed device versus a lower collector –emitter

breakdown

voltage

.In addition,a range of other de-vices and passive components are also available.The circuits are processed on a

300-m-thick p-type substrate that can be thinned to

180m with a conductivity of 0.1S/m.

Fig.9.Conceptual schematic of how the absorptive SPST is used to produce RF envelope pulses at a carrier frequency of 24.15GHz.

A.Generation of 1-ns and Sub-1-ns RF Pulses at a Carrier Frequency of 24GHz

The transmitter is required to generate RF pulses between approximately 500ps and several nanoseconds in length at a carrier frequency of approximately 24GHz.The pulse length is defined as the width of the carrier envelope that is transmitted by the sensor.A brief summary of operation follows,but more in-depth information regarding this design may be found in [33].The pulse-former consists of an absorptive single-pole single-throw (SPST)switch that gates a CW oscillator for a defined amount of time either toward the Tx antenna or,as is more usual,because of the very low mark space ratio

(0.4%)that the power spectral limitations require,into the absorptive load.The trigger that both turns the pulse former on and controls the length of the pulse is controlled by an integral switch driver that allows for the very fast switching required.A conceptual schematic illustrating the operation of the circuit is shown in Fig.9.

The -parameters of the switch in CW operation were measured on-wafer using a vector network analyzer.The input reflection coefficient and the forward transmission coefficient are presented in Fig.10.The switch exhibits an excellent input and output match in either state —transmission or ab-sorptive —and provides transmission gain for the input signal between 14.2–25.5GHz.The switch isolation is better than 30dB over the 15–35GHz frequency range.However,given that the purpose of the switch is to generate very fast UWB pulses,it is far more illustrative to measure the operation of the switch in the transient/time domain.The switch was fed with a 0-dBm 24.15-GHz signal from an external synthesizer —again,on-wafer —and a variable baseband differential control signal fed to the switch driver so that the pulse duration could be adjusted.The output spectrum of the pulse-former when the control signals were set to produce a 1-ns-long pulse envelope with a 30-MHz PRF is illustrated in Fig.6.The spectrum was measured by connecting the output of the switch directly to a spectrum analyzer set with a 5-GHz span from 21.5to

26.5GHz.The

expected

function can be clearly seen with the first null appearing at a frequency offset of 1GHz corresponding to a 1-ns pulse.The 10-dB points as specified in the details of the FCC ruling here indicate a spectral occu-pancy of 1.4GHz.Discrete spectral lines at the pulse repetition frequency (PRF)can also be discerned within the spectral https://www.wendangku.net/doc/3f19058144.html,stly,the LO carrier breakthrough can be clearly seen.Measurement of the output pulse train directly in the time

domain is difficult due to bandwidth limitations with measuring equipment,but is possible by phase locking a microwave transition analyzer to both the CW synthesizer and baseband control signal.Fig.11illustrates the measured envelope of a 1-ns-long pulses at 24.15GHz.Pulses of length of up to 5-ns long were measured.Rise-and fall-transition times for pulses of approximately 60ps are indicated in this figure.B.Tx Antenna Driver Amplifier

The maximum required output power of the Tx antenna driver amplifier may be determined from considering the maximum regulated EIRP spectral density,the requirement on antenna coverage,and the resultant antenna gain.The loss of intercon-nects between the amplifier IC,its package,and the feed lines to the antenna should be also considered in this calculation.These have been extracted for our application by a combination of measurement and full-wave electromagnetic (EM)simulation.For the sensor specification detailed in Table III,these factors result in a peak CW output power requirement for the antenna driver stage from

approximately 10

to 12dBm at the 1-dB

compression

point

.This figure assumes a 500-ps pulse whose peak-power spectral density is distributed equally across the frequency band.While unrealistic,this assumption is the easiest way to calculate the required amplifier power.Another goal is to ensure that the driver amplifier is either close to,or in,compression.This is advantageous in a power-limited system such as the SRR sensor to ensure that maximum performance is possible over variations in temperature and unit to unit.

A microphotograph of the amplifier stage is shown in Fig.12.The amplifier is comprised of three common-emitter (CE)class-A stages.The final output stage is realized with multiple transistors in parallel to ensure sufficient output power.The potential output power of this stage can be adjusted by increasing the dc-bias current.Measured on-wafer results

for

the

of the amplifiers ’output stage at 24GHz as a function of bias current is shown in Fig.13.For a bias current of 40mA,the amplifier begins to compress

at 10dBm.Increasing the dc-bias current of the amplifier output stage to 60mA can increase this to

over 12dBm.This is an impor-tant tradeoff in the design process.

The 14-dBm operating environment of SRR around the vehicle is extremely harsh,with no good thermal ground path,high ambient temperatures,and an uncertain degree of airflow.Increasing the thermal power dissipation of the sensor is not a decision to be taken lightly.Higher efficiency could be obtained from the amplifier by modifying the class of the design to class AB or higher at the output stage.However,the conclusion stands in that the required levels of output power required for the antenna driver amplifier are easily available without requiring an unacceptably large device/circuit or excessive current consumption.C.I/Q Homodyne Down-Conversion Rx IC

An overview of the operation of the Rx chain has already been given in Section III.The schematic of the IC shown in Fig.14represents the receiver from the antenna input to the output of the baseband amplifier.As always,the design of the receiver is a balance of minimizing the cascaded noise figure,increasing

Fig.10.Measured reflection coefficient (S 11)and forward transmission coefficient (S 21)S -parameters of the SPST switch in the Tx (ON )and absorptive (OFF )

states.

Fig.11.Measured envelope of the 24-GHz RF pulse signal at the output of the SPST switch.The differential control signal was set to produce a 1-ns pulse.The corresponding frequency spectrum can be seen in Fig.6for a pulse train with a 30-MHz

PRF.

Fig.12.Microphotograph of the Tx antenna driving amplifier.The parallel transistor cells that form the output stage may be clearly seen.The complete amplifier,including bias circuitry,is 800 m 2500 m.

the dynamic range by maximizing the input compression power,and allocating an appropriate gain budget through the receiver chain.

The LNA is common to both I and Q channels,and has a three-stage lineup.The first two stages are source matched

to

Fig.13.Gain characteristic of the output stage of the Tx antenna driver amplifier.The saturation point of the amplifier is shown to vary as a function of the bias

current.

Fig.14.Schematic diagram of the I/Q homodyne down-conversion IC.

minimize the added noise from the amplifier,and the third stage is included to increase the gain of the amplifier.Careful study of the source matching circuit for the first stage is required.It is possible that the loss of the matching elements used —on an Si substrate at 24GHz —may result in a worse overall noise figure than a similar amplifier where only minimal matching is added.Similar concepts have been used in the design of the low-noise mixer [44].

The mixers are both designed to have a conversion gain that is marginally higher than their noise figure.Study of the cas-caded noise figure of the receiver indicates that once the gain of the mixer is larger than the noise figure,no further benefit is possible.Although an increase in the conversion gain of the device is possible,it is at the cost of higher power consump-tion [44].The mixer is the dominant limitation in

the

of the receiver,given that it has an amplifier preceding it.Satura-tion for the receiver per se is not usually an issue,but two in-stances must be borne in mind.The first is the recovery time of the receiver from saturation.A common problem with the in-stallation of many sensors behind the same fascia is transmitted power being reflected directly back into the receiver.This will occur with every Tx pulse,regardless of the rang gate under in-spection,momentarily blinding the sensor to reflected power.As the sensor sweeps the range gates,the receiver must recover in a sufficient time to begin receiving information.The second issue is one of dynamic range and occurs when a highly reflec-tive target may be close to a much smaller one.In this case,re-ceiver saturation may mean that the smaller target is missed.For this reason,it is important to have a variable gain IF amplifier in the Rx chain.This is needed because,although the transient characteristics of the downconverter itself may be adequate,fur-ther stages of the receiver with lower frequency filters may have transient behaviors that prevent them from recovering in time.The first integration and detection stages along with the dc offset sampling and correction circuits are realized in the instance of this particular IC in a separate ASIC.

The IC was manufactured using the Atmel SiGe2rf process.

The receiver IC measures 1.3

mm

1.7mm in size.This includes the LNA,power splitter,and dual-channel mixers and IF stages.Fig.15illustrates the measured on-wafer perfor-mance of the IC.An external frequency synthesizer was used to supply a 24-GHz LO signal with a power level of 0dBm.The single-channel conversion gain and noise-figure characteristics of the receiver chip were measured directly using the down-converter option of an Agilent N8975A noise-figure analyzer with a variable IF.The measurement was performed on either side of the 24-GHz LO frequency in order to extract both upper and lower sideband information.The conversion gain includes the contribution of the LNA,power splitter,mixer,and baseband buffer amplifier.The noise figure is similarly given for the entire cascade.The conversion gain varies as a function of the IF bandwidth.For an IF of 1GHz —both upper and lower sideband —the gain is approximately 38dB,and slowly increases as the IF frequency is decreased to a peak value of approximately 44dB.Separate measurements on both the LNA and mixer [44]indicate that most of this gain slope can be attributed to the IF baseband amplifier.This is principally due to the amplifier being unmatched due to

the

Fig.15.Measured on-wafer single-channel Rx gain and noise figure for RF frequencies between 23–25GHz.The LO frequency is held constant at 24GHz.

need for dc coupling.Some resistive matching on the amplifier could improve this at the expense of a reduction in gain.The measured gain value also compares well with the gain budget through the Rx

chip:18-dB gain for the LNA,6-dB loss in the passive

splitter,8-dB conversion gain in the mixer,

and 20dB in the IF amplifier.The noise figure,also shown in Fig.15,is symmetrical around the carrier frequency and is approximately 6.5dB across the entire IF bandwidth.This figure is measured on-chip and does not include the loss of the package transitions and board interconnects.However,it is in line with the required system level value of 8dB to achieve the detection of the minimum RCS target at the maximum range with the required probability of detection.D.Planar Antenna With Low-Elevation Sidelobes

The challenging specifications imposed by the FCC ruling on sidelobe levels in elevation are one of the principal consid-erations in antenna design.Typical Tx antenna requirements of each sensor in the network are 60azimuth beamwidth,12–15elevation beamwidth,and 11–14-dBi gain depending upon the specific implementation.The eventual elevation sidelobe level is,therefore,a function of antenna gain and the particular wave-form selected.The FCC specification only stipulates the emis-sion level of the radiated power spectral density and is,there-fore,flexible,relative to the conducted power level.For our transmitter,this translates to a maximum antenna sidelobe level

in elevation

of

25dB relative to bore sight

at 30.The most appropriate type of planar antenna for this applica-tion is a microstrip patch antenna.The microstrip patches can be direct fed or slot fed.The slot or aperture coupled patch an-tennas are preferred in order to achieve very low sidelobes [45],[46].With this type of approach,the feed network is completely isolated from the radiating elements by the ground plane.This avoids degradation of sidelobes and cross-polarization due to radiation from the feed.In addition,the aperture coupled patch-antenna configurations have broader frequency bandwidth and provide an additional degree of freedom in choosing substrates for the feed network and radiating elements.The construction

Fig.16.Cross-sectional view of the slot-coupled patch

array.

Fig.17.Modeled versus measured results for an FCC specification compliant antenna with 025-dBc elevation sidelobes.

of the slot coupled patch array is shown in Fig.16.There is an air cavity behind the slots to prevent back radiation.This improves the gain and front-to-back radiation ratio.To obtain very low sidelobes,a steep tapering of the element amplitudes is required in addition to precise phase balance.Very low power at the end patches requires very narrow

(100m)microstrip lines.Full-wave simulation tools were used to separately opti-mize both the feed network and radiation patterns.The simu-lated and measured radiation patterns are shown in Fig.17.The measured performance is comparable to the simulated perfor-mance:a measured sidelobe level of 25.5dB versus a com-puted value of 28dB,and a measured 3-dB beamwidth of 17.4versus a computed value of 17.5.However,it was ob-served in testing that the sidelobes are very sensitive to the as-sembly of the SRR antenna module,and care in assembly pro-cedures is needed to ensure a high yield.

VI.C ONCLUSIONS AND C OMMENTS

The last few years has seen a reinvention of the way that microwave and millimeter-wave systems are thought about and designed.The increasing commercialization of high-frequency consumer products has been furthered by rapid advancements

in semiconductor technology,improved packaging and as-sembly materials and techniques,and increased flexibility in processing capability through field-programmable gate array (FPGA)processors.Improved predictive modeling and simulation tools have meant that products that were not feasible just ten years ago are now either already commonplace or just over the horizon.The concept of a smart vehicle capable of making decisions to aid,or ultimately intervene,in the driving process has long been a popular detail in science fiction,yet developments are under way to bring it to reality.

The UWB ruling by the FCC that permits the usage of SRR sensors has been critical to this process of development.Range resolution of the accuracy required for almost all of the short-range sensing requirements in vehicular applications requires considerable bandwidth allocation that is unimaginable at power levels above the standard EIRP level for spurious emissions.The uncertainty of how large numbers of radiators will affect other spectrum users by lifting the general level of background noise is a genuine cause for concern,and excuses a degree of tentative-ness in regulatory agencies in passing legislation that permits such usage.However,the importance of reducing automotive fa-talities and injury rates is a strong motivation for both regulators and industry.The phased-in introduction of automotive radar sensors —whether in the U.S.or worldwide —will permit a long opportunity to study the effects and issues associated with the multiuser spectral occupancy that the UWB regulations permit,and the effect on automotive occupant safety.

Of course,the fact that this product introduction is occurring at all is due,in part,to the convergence of what are key enabling technologies.Examples of some of the more critical of those technologies —millimeter-wave ICs realized with low-cost semiconductors,the ability to package the same ICs in stan-dard plastic packages,low-loss PCBs that facilitate efficient antennas —have been presented in this paper.The rapidity with which these technologies further advance is keenly anticipated.

A CKNOWLEDGMENT

The authors extend many thanks to K.Puglia,https://www.wendangku.net/doc/3f19058144.html,bitt,J.Schaefer,J.Ruel, C.Luck,N.Mohler,M.Paradie,M.Marple,H.Henftling,and D.Klotzbuecher,all of the Automotive Group,M/A-COM,Lowell,MA,for their help and support in this development.

R EFERENCES

[1]“First report and order,revision of part 15of the commission ’s rules

regarding ultra wideband transmission systems,”FCC,Washington,DC,ET Docket 98-153,2002.

[2]R.Aiello and G.D.Rogerson,“Ultra-wideband wireless systems,”IEEE

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1986.

Ian Gresham(M’90–SM’03)received the B.Eng

and Ph.D.degrees in electronic and electrical

engineering from The University of Leeds,U.K.in

1988and1994,respectively.

From1988to1990,he was with Marconi

Command and Control Systems,Leicester,U.K.,

where he was a Microwave Engineer in the fields

of fire-control and Naval radar.He then joined the

Microwave and Solid State Research Group,The

University of Leeds,where his research focused

on low phase-noise signal generation at microwave and millimeter-wave frequencies.Upon graduation,he became a Senior Engineer with M/A-COM,Milton Keynes,U.K.,where he designed microwave components and sub-systems for both military and commercial applications, prior to concentrating on HBT and pseudomorphic high electron-mobility transistor(pHEMT)-based MMIC design for the wireless industry.In1998, he joined Corporate Research and Development,M/A-COM,Lowell,MA,as a Principal and then Senior Principal Engineer.Since that time,his research interests have been oriented toward the commercialization of millimeter-wave systems with particular emphasis on automotive24-GHz short-range sensors and77-GHz ACC radars.

Alan Jenkins(M’02)was born in Southampton,U.K.,in1967.He received the Bachelors and Masters degree from Bradford University,Bradford,U.K.,in 1989and1990,respectively,and the Ph.D.degree in microwave applications of high-temperature superconductors from Oxford University,Oxford,U.K.,in 1995.

From1990to1991,he was a MMIC Designer with the then GEC Marconi Caswell Research Center,where he designed GaAs HBT circuits.In1991,he joined Oxford University.From1994to2000,he was initially a Junior Re-search Fellow and then a Senior Researcher with the Communications Group, Oxford University,where he investigated numerous aspects of thallium-based superconductors for commercial microwave applications,propagation studies, and imaging techniques for third-generation(3G)cellular applications.During this time,he was also Director of S-comm,a successful technical consulting company.In March2001,he joined Corporate Research and Development, M/A-COM,Lowell,MA,where he has been active in the design of SiGe analog and digital circuits at millimeter-wave frequencies for large-volume applications.He has authored or coauthored over42technical papers.He holds three patents on various topics in microwave and millimeter-wave engineering. Robert Egri(S’78–M’79)received the Ph.D.degree from The Johns Hopkins University,Baltimore,MD,in1979.

In1992,he joined M/A-COM,Lowell,MA,as a System Engineer,where he has been involved with communications and radar-related problems.Prior to

joining M/A-COM,he was with Raytheon and COMSAT

Laboratories.

Channabasappa Eswarappa(M’93–SM’97)

received the M.Tech.degree in electrical engineering

from the Indian Institute of Technology,Kanpur,

India,in1983,and the Ph.D.degree from the

University of Ottawa,Ottawa,ON,Canada,in1990.

From1990to1992,he was with the Indian

Telephone Industries,Bangalore,India,where he

developed microwave circuits for telecommunication

and satellite networks.From1992to1997,he was

a Post-Doctoral Fellow and then a Senior Research

Engineer with the Department of Electrical and Computer Engineering,University of Victoria,Victoria,BC,Canada.His research focused on the development of time-and frequency-domain numerical techniques for modeling EM fields and computer-aided design of microwave and millimeter-wave circuits,antennas,and radar cross-sections.In1997, he joined M/A-COM,Lowell,MA,where he is currently a Senior Principal Engineer.He has been involved with various projects,such as development of compact low-loss filters for wireless communications base stations,power combiners for a28-GHz local multipoint distribution system(LMDS),mil-limeter-wave plastic packages,and antennas for24-and77-GHz automotive radars.He has authored or coauthored over60technical publications in referred

journals and conference proceedings.He holds four

patents.

Noyan Kinayman(S’94–M’96)was born in Ankara,

Turkey,in1968.He received the B.Sc.and M.Sc.

degrees from the Middle East Technical University,

Ankara,Turkey,in1990and1993,respectively,and

the Ph.D.degree from Bilkent University,Ankara,

Turkey,in1997,all in electrical engineering.

From1990to1994,he was an Electrical En-

gineer with Aselsan Inc.(a military electronics

company),Ankara,Turkey,where he developed

computer-controlled test stations to expedite testing

of military communication equipments.From1994 to1997,he was a Research Assistant with the Electromagnetics Group,Bilkent University,where he studied novel algorithms to simulate printed circuits. Upon completion of his doctoral studies,he joined Corporate Research and Development,M/A-COM,Lowell,MA,as Senior Electrical Engineer.He is currently with the same department as a Principal Electrical Engineer.His main responsibilities are EM analysis,modeling,and microwave circuit design. He has authored and coauthored21technical publications in peer-reviewed international journals and conferences.He holds two patents.He has developed commercially available full-wave EM simulation software to simulate planar microstrip circuits.His main professional interests are EM theory,numerical solution of EM problems,and model extraction of passive microwave printed

circuits.

Nitin Jain(S’90–M’91)received the B.Tech.

degree in electronics from the Indian Institute of

Technology,Madras,India,in1986,and the M.S.

and Ph.D.degrees in solid-state electronics from

Rensselaer Polytechnic Institute,Troy,NY,in1989

and1991,respectively.His doctoral research focused

on MESFET,pHEMT,and diode–gate MESFET

control devices.

From1991to1998,he was with Corporate Re-

search and Development,M/A-COM,Lowell,MA,

where he was a Senior Principal Engineer.His re-search focused on the commercial use of EM simulators,passive and active circuit modeling,multilayer PCB modeling,high-power p-i-n switches,mil-limeter-wave circuits,and millimeter-wave module development.He has been involved with various commercial millimeter-wave projects,including the de-sign and production of an LMDS receiver at28GHz and the design of a24-GHz collision-avoidance radar.He was also the technical lead of a team that designed a single substrate radar module for a77-GHz ACC application.From1998to 2000,he was with the Electrical and Computer Engineering Department,In-dian Institute of Science,Bangalore,India,as an Assistant Professor.Since November2000,he has been with Anokiwave Inc.,San Diego,CA,where his interests include millimeter-wave circuits and millimeter-wave communication modules,cellular power amplifiers and front-end integration,device modeling, and integrated passive circuits.He has authored or coauthored over25publica-tions in international conferences and journals.He holds11U.S.patents. Dr.Jain was the recipient of the1990IEEE Graduate Student Fellowship Award.

Richard Anderson received the Ph.D.degree in materials engineering from the Rensselaer Polytechnic Institute,Troy,NY,where he investigated polaron hopping in transition metal–oxide glasses.

He is currently a Senior Principal Engineer with corporate research and de-velopment,M/A-COM,Lowell,MA,where he develops high-frequency pack-aging and assembly processes.Prior to joining M/A-COM,he was involved with process development with Motorola,the Foxboro Company,and3M.He has au-thored or coauthored over30publications in the materials processing for elec-tronics field.He holds eight patents.

Dr.Anderson is a member of the International Microelectronics and Packaging Society(IMAPS)and the Surface Mount Technology Association (SMTA).

Frank Kolak,photograph and biography not available at time of

publication.

Ratana Wohlert received the B.S.E.E.degree from

Southeastern Massachusetts University,North Dart-

mouth,in1984,and the M.S.E.E.degree from North-

eastern University,Boston,MA,in1988.

In1984,she entered the microwave industry.

She is currently with Corporate Research and

Development,M/A-COM,Lowell,MA.Her ex-

periences range from playing an instrumental role

in the pioneering and commercializing of GaAs

MMIC control devices to the design of active

millimeter-wave products.Her design experience ranges to include many MMIC power and LNAs employing HEMT,PsHEMT, and HBT GaAs technology.Her design frequency experience ranges from C-to Q-band.A sample of her publications included35-GHz pulsed HBT MMIC amplifiers,Ka-band GaAs HBT p-i-n diode switches and phase shifters,and Ka-band GaAs HBT LNAs.

Shawn P.Bawell(M’02)received the B.S.degree

in electrical engineering from the Pennsylvania State

University,University Park,in1995,and is currently

working toward the M.S.degree in electrical engi-

neering at the University of Massachusetts,Lowell.

Upon receiving the B.S.degree,he joined

M/A-COM,Lowell,MA,where he was initially

involved with passive and active IC design.His latest

research includes both high-speed digital designs

and high-speed analog design for radar applications

on SiGe technology.

Jacqueline Bennett(S’80–M’81),photograph and biography not available at time of publication.Jean-Pierre Lanteri(M’82)received the M.S.degree in electronics from the Ecole Centrale de Lyon,Lyon,France,in1978,and the Ph.D.degree in semicon-ductor electronics from the University P.Sabatier,Toulouse,Toulouse,France, in1980.

Until1983,he was with Thomson-CSF,Corbeville,France,where he devel-oped GaAs characterization and modeling techniques and tools.In1985,he joined M/A-COM,Lowell,MA,where he has been involved with the imple-mentation of device modeling,statistical process control,automatic on-wafer test,and reliability analysis in its GaAs foundry.Since1990,he has established automated assembly and test facilities producing,in volume,RF integrated cir-cuits(RFICs),Tx/Rx modules,77-GHz cruise-control sensors,and complete private mobile radios.He is currently the Director of Technology for M/A-COM, where he leads its research and development efforts in system-on-chip for mi-crowave/millimeter-wave applications,broad-band switch ICs and packages, and digital linearization of amplifiers.

最新多普勒雷达系统仿真

多普勒雷达系统仿真

精品好文档，推荐学习交流 摘要 现代通信系统要求通信距离远、通信容量大、传输质量好，作为其关键技术之一的调制解调技术一直是人们研究的一个重要方向。本文以MATLAB为软件平台，充分利用其提供的通信工具箱和信号处理工具箱中的模块，对数字调制解调系统进行Simulink设计仿真，并且进行误差分析。 数字化正交数字化正交调制与解调是通信系统中十分重要的一个环节，针对不同的信道环境选择不同的数字化正交数字化正交调制与解调方式可以有效地提高通信系统中的频带利用率，改善接收信号的误码率。本设计运用Simulink仿真软件对二进制调制解调系统进行模型构建、系统设计、仿真演示、结果显示、误差分析以及综合性能分析，重点对BASK，BFSK，BPSK进行性能比较和误差分析。在实际应用中，视情况选择最佳的调制方式。 本文首先介绍了课题研究的背景，然后介绍系统设计所用的Simulink仿真软件，随后介绍了载波数字调制系统的原理，并根据原理构建仿真模型，进行数字调制系统仿真，最后对设计进行总结，并归纳了Simulink软件使用中需要注意的事项。本文的主要目的是对Simulink的学习和对数字调制解调理论的掌握和深化，为今后在通信领域继续学习和研究打下坚实的基础。 关键字：排通信系统，Simulink仿真，数字化调制解调，BASK，BFSK

精品好文档，推荐学习交流 ABSTRACT TheThe Modern communication systems require communication distance, large communication capacity, good transmission quality, as one of its key technologies modem technology has been an important direction for researchers. In this paper, MATLAB software platform, providing full use of its communications toolbox and signal processing toolbox module, digital modulation and demodulation system Simulink design simulation and error analysis. Modulation and demodulation is a very important part of the communication system, for different channel environment to select different modulation and demodulation system can effectively improve the spectrum efficiency in a communication system, improve the bit error rate of the received signal. This design using Simulink simulation software binary modulation and demodulation system modeling, system design, simulation demo showed that the error analysis and comprehensive performance analysis, focusing on the BASK, BFSK, BPSK performance comparison and error analysis. In practice, as the case may select the best modulation. This paper describes the background of the research, then describes the system design using Simulink simulation software, then introduced the carrier digital modulation system of principles, and build a simulation model based on the principle of digital modulation system simulation, and finally the design summary and induction Simulink software matters that need attention. The main purpose of this paper is to study and Simulink digital modem theory of mastery and deepening for the future to continue learning and research in the field of communication and lay a solid foundation. Key Words: queuing theory, demand management, telecom offices

2019年时政：2月21日国内篇

[键入文字] 2019 年时政：2 月21 日国内篇 1、脱贫攻坚三年行动开局良好我国农村贫困发生率降至1.7% 记者从国新办吹风会上获悉：2018 年我国减少农村贫困人口1386 万，连续6 年超额完成千万减贫任务，贫困发生率下降到1.7%，全国已有153 个贫困县宣布脱贫摘帽，2018 年预计还有284 个贫困县退出。 2、长江经济带环境污染犯罪可从重处罚 随着环境污染犯罪案件多发高发，执法司法机关普遍反映实践中存在着确定管辖 难、调查取证难、司法鉴定难、法律适用难等突出问题，亟待通过顶层设计予以解 决。 20 日，最高人民法院、最高人民检察院、公安部、司法部、生态环境部首次就办理环境污染刑事案件有关问题联合出台《关于办理环境污染刑事案件有关问题座谈会纪要》，要求统一法律适用和政策把握，依法准确有效惩治环境污染犯罪，形成各部门依法惩治环境污染犯罪的合力。 此次《纪要》也对长江经济带区域污染环境犯罪规定了相关从重处罚的情形，规定 对于发生在长江经济带11 省（直辖市）的跨省（直辖市）排放、倾倒、处置有放射性的废物、含传染病病原体的废物、有毒物质或者其他有害物质的，将从重处罚。 3、农业农村部要求做好斑海豹幼崽救护工作 近日，大连破获一起特大盗捕斑海豹案件。案件发生后，农业农村部要求严查此事 并务必做好斑海豹幼崽救护工作，同时派员会同辽宁省农业农村厅赶赴大连，就案件 处理和斑海豹救治情况进行指导和对接。 下一步，农业农村部还将指导协调地方渔业渔政部门，妥善做好斑海豹幼崽收容救 助工作，待专家和兽医联合评估认可具有野外生存能力后再分批放归野外。农业农村 部还将进一步加强海洋馆等繁育场所规范管理，切断可能的非法贸易链条，联合有关 1

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第28卷第5期 2004年9月大气科学Chinese Journal of Atmospheric Sciences Vol 128　No 15Sept.2004 2003205208收到,2003210214收到修改稿 3中国科学院百人计划和上海市光科技计划共同资助 基于斐索干涉仪的直接探测多普勒测风激光雷达 3刘继桥　陈卫标　胡企铨 (中国科学院上海光学精密机械研究所先进激光技术与应用系统实验室,上海201800) 摘　要 提出结合多光束斐索(Fizeau )干涉仪和CCD 探测器的条纹图像技术,测量地球边界层下的三维风场的直接探测多普勒激光雷达技术。在分析Fizeau 干涉仪的物理特性和光谱特性以及影响测量多普勒频移的因数和改进方法的基础上,提出一套切合实际的直接探测多普勒激光雷达系统参数。并利用该参数进行性能评估分析,模拟不同干涉仪参数对风速精度的影响,得出一个优化的干涉仪物理参数。模拟结果显示,系统可以获得小于1m s -1的水平风速精度。这些分析,为建立实际的激光雷达系统提供设计依据。 关键词:多光束斐索干涉仪;直接探测;多普勒激光雷达;风速 文章编号　100629895(2004)0520762209 中图分类号　P415 文献标识码　A 1　引言 大气风场是各种天气过程、大气化学成分循环和海气相互作用的主要动力,因此大气风场探测在气象、环境等领域中有着极其重要的地位。多普勒激光雷达已经被认为是精确测量三维风场的有效手段[1]。从全球风场的测量来看,直接探测多普勒激光雷达技术相对相干技术来说存在一定的优势[2]。边缘检测[3]和条纹图像[4]是目前直接探测多普勒激光雷达中最主要的两种多普勒频移测量技术。边缘检测常采用高分辨率的法—伯(FP )干涉仪[3]或者分子[5]、原子吸收线的翼作为鉴频器,其测量灵敏度依赖于分子和气溶胶的后向散射比和风速大小;条纹图像技术则是利用干涉条纹的移动直接测量多普勒频移。Mc G ill 等[6]详细分析、比较了两种测量技术,认为两种技术在风速测量精度十分接近。Mc Kay 等[7]从星载系统的角度比较两种技术,认为条纹图像技术更适合于研制星载激光雷达系统。 最初的条纹图像技术采用FP 干涉仪和图像光电探测器(IPD )得以实现,但这种多阳极光电倍增管的量子效率比较低,而且像元数很有限[8]。Irang 等[9]演示了利用CCD 探测器的条纹图像的直接探测激光雷达,系统利用复杂的二元光学技术将环形条纹转换成点阵[10],增加系统复杂性。因此,相关学者把目光转移到寻找更加适合的干涉仪来代替FP ,如M 2Z 干涉仪[11]和Fizeau 干涉仪[12]。Mc Kay [12]首次分析了利用Fizeau 干涉仪进行多普勒频移检测,其分析是较初步的,也没有针对具体系统进行分析。由于Fizeau 干涉形成的是直线条纹,这样可以利用量子效率较高的线阵固体探测

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第9卷第21期 2009年11月1671 1819(2009)21 6532 04 科学技术与工程 ScienceTechnologyandEngineering 2009 Sci Tech Engng 9 No 21 Nov.2009 Vol 通信技术 半导体激光器驱动电路设计 何成林 (中国空空导弹研究院,洛阳471009) 摘要半导体激光驱动电路是激光引信的重要组成部分。根据半导体激光器特点,指出设计驱动电路时应当注意的问题,并设计了一款低功耗、小体积的驱动电路。通过仿真和试验证明该电路能够满足设计需求,对类似电路设计有很好的借鉴作用。 关键词激光引信半导体激光器窄脉冲中图法分类号 TN242; 文献标志码 A 激光引信大部分采用主动探测式引信,主要由发射系统和接收系统组成。发射系统产生一定频率和能量的激光向弹轴周围辐射红外激光能量,而接收系统接收处理探测目标漫反射返回的激光信号,而后通过信号处理系统,最终给出满足最佳引爆输出信号。由此可见,激光引信的探测识别性能很大程度上取决于激光发射系统的总体性能,即发射激光脉冲质量。而光脉冲质量取决于激光器脉冲驱动电路的质量。因此,半导体激光器驱动电路设计是激光引信探测中十分重要的关键技术。 图1 驱动电路模型 放电,从而达到驱动激光器的目的。 由于激光引信为达到一定的探测性能,通常会要求激光脉冲脉宽窄,上升沿快,一般都是十几纳秒甚至几纳秒的时间。因此在选择开关器件时要求器件开关速度快。同时,由于激光器阈值电流、工作电流大 [1] 1 脉冲半导体激光器驱动电路模型分析 激光器驱动电路一般由时序产生电路、激励脉冲产生电路、开关器件和充电元件几个部分组成,如图1。 图1中,时序产生电路生成驱动所需时序信号,一般为周期信号。脉冲产生电路以时序信号为输入条件。根据其上升或下降沿生成能够打开开关器件的正激励脉冲或负激励脉冲。开关器件大体有三种选择:双极型高频大功率晶体管、晶体闸流管电路和场效应管。当激励脉冲到来时,开关器件导通,